Keithley 427 User manual

INSTRUCTION MANUAL
MODEL 427
CURRENTAMPLIFIER
0 1975, KEITHLEY INSTRUMENTS, INC.
CLEVELAND, OHIO, U.S.A.
DOCUMENTNUMBER29103

CONTENTS MODEL 427
CONTENTS
1. GF,Ng3,&.L DESCRIPT~ON-------------------------------.-------- 1
2. OPERATION-------------------------------------------------- 4
3. APPLICATIONS----------------------------------------------- 9
4. ACCESSORIES------------------------------------------------ 10
5. ClRC"IT DESCRIPTION---------------------------------------- 12
6. REp~Cp‘&LE PARTS------------------------------------------ 17
7. CALIBRATION------------------------------------------------ 26
SCNEMATICS---------------------------------------------------- 33
1074

MODEL 427 ILLUSTRATIONS
ILLUSTRATIONS
Figure No. Title Page
1 Front panel. __-___---___-_____-___________________ 1
2 Front pane1 Controls. ________--______-____________ 3
3 Rear Panel Controls. - 3
4 shunt Method Measurement, -----------------__------ 4
5 Effect of Shunt capacitance. ---------------------- 4
6 Compensation for Shunt Capacitance. --------------- 5
6b Extended Frequency Response. ---------------------- 5
7 Fee&a& Method. _--__-_-__________-_______________ 6
8 Input Voltage N&se. -------------___-------------- 6
9 Bandwidrh of Feedback System. --------------------- 7
9b Effect of filter on Noise spectrum. -------------- 7
9, Effect of Input Capacitance on Noise. ------------- 7
10 Frequency Compensafian. ---------_____--_-__------- 7
11 Plot of Noise-Improvement Contours. __- _____ -__-__ 9
12 Block Diagram of a High-Speed Current Amplifier, -- 12
13 Filter circuit. ------_______--_-__-_______________ 13
14 power s”pp~y Regulator. --------------------------- 13
15 current suppression. ------------__---------------- 13
lb COmpOnent Layout - PC-291. ------------------------ 13
17 component Layout - pc-290, -----------------___---- 14
18 component Layout - pc-2?39. ---------_------___-____ 15
19 COmpOnenf Layout - P&292. ---------------__------ 15
20 chassis - Top vie”. ------------___--------------- 16
21 Chassis Assembly - Exploded “few. ----------------- 19
22 Bottom CO”er Assembly. -------------------------- 19
23 Measurement of Input Voltage Drop. ---------------- 27
24 Measurement of Rise Time. -------_--_____-___-_____ 28
25 Measurement of Filter Rise Time. ------------------ 29
1074 iii

SPECIFICATIONS MODEL 427
SPECIFICATIONS
RANGE: 10’ to 10” volts/ampere in eight decade ranges.
(10-13 ampere resolution to 10.” ampere full output).
OUTPUT: *10 volts at up to 3 milliamperes.
OUTPUT RESISTANCE: Less than 10 ohms dc to 30 kHz.
OUTPUT ACCURACY: 12% of reading to the lo9 vattsl
ampere range, +4% of reading on the 1O’O and 10”
volts/ampere ranges exclusive of noise. drift and current
offset.
RISE TIME (10% to 90%): Adjustable in lx and 3.3x steps
from “Fast Rise Time” listed below to 330 rnsec.
NOISE VS. RISE TIME’:
I FASTRISETlML
STABILITY: Current offset doubles per 10°C above 25°C.
Voltage drift is less than 0.005% per “C and less than
0.005% ,,er da” of full outDut after l.hour warmur,.
,-
OFFSET CURRENT: Less than lQLz am&e at 25”C’and up
to 7001, relative humidity.
CURRENT SUPPRESSION: lo-lo ampere to 10-a ampere in
eight decade ranges with 0.1% resolution (lO.turn poten.
tiometer). Stability is +O.Zo/. of suppressed value per “C
bO.Z% per day.
INPUT VOLTAGE DROP: Less than 400 /.IV for fullaxle
output on the 10” to 10” volts/ampere ranges when
properly zeroed.
EFFECTIVE INPUT RESISTANCE: Less than 15 ohms on the
10” and 105 volts/ampere ranges. increasing to less than
4 megohms on the 10” volts/ampere range.
MAXIMUM INPUT OVERLOAD: Transient: 1000 volts on any
range for up to 3 seconds using a Keithley (or other 10
mA-limited) highaoltags supply. Continuous: 500 volts
on the 10” to 1O’voltsjampere ranges, decreasing to ‘ZOO
an the 10”. 70 on the lo5 and 20 volts on the 10’ volts,
ampere ranges.
OVERLOAD INDICATION: Lamp indicates pre-filter or post.
filter overload.
DYNAMIC RESERVE: 10 (20 dB).
CONNECTORS: Input: (Front) ENC. Output: (Front and
Rear) BNC.
POWER: 90.125 or 180.250volts (switch selected), 50.60 Hr.
5 watts.
DIMENSIONS; WEIGHT: Style M 3%” half.rack, overall bench
size4’~highx8%‘widex12L/a”deep(100x217 x310 mm).
Net weight, 7 Ibs. (3.0 kg).
i” 1074

MODEL 427 GENERAL DESCRIPTION
SECTION 1. GENERAL DESCRIPTION
l-1. GENERAL; The Model 427 Current Amplifier is a C. Built-in Current Suppression. Small changes in
high-speed, feedback-type amplifier with particular the signal level can be measured since large ambient
features useful for automated semiconductor testing, current levels can be easily suppressed.
mass spectrometry, and gas chromatography applications. d. Overload Indication. Accurate measurements are
assured since overloads are automatically indicated.
l-2. FEATURES. e. Variable Cain. The GAIN Switch is designated
a. Wide Dynamic Range. Selectable rise times permit in eight gain positions from lo4 to 1011 volts per
low-noise operation important when resolving small ampere - therefore gain adjustment is straight forward.
current levels. f. Variable Kise Time. Optimum response can be
b. High Speed. Typical resolution is 20 picoamperes
out of a 10sSZpere signal with a 100 microsecond selected for each gain setting since a separate RISE
TIME switch is provided on the front panel.
rise time.
0471 1

GENERAL DESCRIPTION MODEL 427
TABLE 1-l.
Front Panel Controls and Terminals
Functional Description Paragraph
PUSH Power Switch (S302)
GAIN Switch (5201)
RISE TIME (5101)
SUPPRESSION
MAX AMPERES Switch (S303)
FINE Control (R333)
POLARITY Switch (5304)
ZERO ADJ Control (R235)
INPUT Receptacle (5202)
OUTPUT Receptacle (5102)
~ OVERLOAD Indicator (DS302)
Controls power to instrument.
sets gain in Volts per ampere.
Sets rise time Fn milliseconds.
Sets maximum suppression.
Adjusts suppression.
Sets polarity of suppression.
Adjusts output zero.
Input source connection.
Output connectFon.
Indicates overload condition.
2-4, al
2-4, a2
2-4, a3
2-4, a4
2-4, a5
2-4, a6
2-4, a7
2-3, a
2-3, b
2-5, d
TABLE 1-2.
Rear Panel Controls and Terminals
Control or Terminal Functional Description Paragraph
Line Switch (5301)
PowerReceptacle (P305)
Fuse (F301)
Sets instrument for 117V or 23411.
Connection to line power.
Type 3AG Slow-Blow, 117V @ l/4 A (w-17)
234V @ l/S A (w-20)
2-4, b
2-3, c
0lJTPuTReceptacle (5103) Output connection. 2-3, b
WARNING
Using a Line Power Cord other than the one supplied
with your instrument may result in an electrical
shock hazard. If the Line Power cord is lost or
damaged, replace only with Keithley Part No. CO-7.
0878

MODEL 427 GENERAL DESCRIPTION
GAIN
SWITC
S20
,----SUPPRESSION-,
FINE POLARITY
ADJUST SWITCH
INPUT
5202
. P
ZERO ADJ OVERIDAD Power OUTFW
R235 DS302 S302 5102
FIG"RF 2. want Panel Controls.
I
FIGURE 3. Rear Panel Controls.
0471

OPERATION
SECTION 2. OPERATION
MODEL 427
2-1.
MEASUREMENT CONSIDERATIONS.
a. Current-Detection Devices. The DleasUrement of
‘small electrical c~rrent8 has been the basis for a
number of instrumental methods used by the analyst.
Ion chambers, high-impedance electrodes, many forms
of ch=“metog=aphic detectors, phototubes and multipli-
ers are coaronly-used t=ansduce=a which eequire the
measurement of small currents. Devices used for this
measurement a=e often called electraaeters.
b.. In any measure-
ment, if e”“=ce noise greatly exceeds that added by
the inst=“mentatian, optimization of instrumenteti””
is unimportant. When source noise approaches theoret-
ical minimum, optimization of instrumentation charac-
teristics becomes imperative. TO determine the cate-
gory into which this meas”=ement falls. the researcher
needs t” be familiar with the characteristics which
impoee theoretical and practical limitations on his
me*surement . Most researchers a=e familiar with the
theoretical limitations present in voltage measueements
The noise inceeases with 8”“=ce realstance, and the
familiar equation for the mean-square noise voltage is
q = 4kTRAf Eq. 1
whe=e k is the Boltzma”” Constant, T ia the absolute
temperature of the s”“=ce resistance R, and Af is the
noise bendwidth( 3 times the 3 dB bandwidth for a
single RC rolloff.) In the case of cureent measure-
ments it Is more appropriate to consider the noise
current generated by the ~l”“=ce and load resistances.
The mean squaee noise c”==ent generated by a resistor
is given by Eq. 2.
FIGURE 4.
In the shunt method c”=re”t is measured by
the voltage drop ac=“ee a resistor.
Prom this equation it is irmnediately apparent that the
m%QB”rement of emall current =equi=es large values of
R, i.e., high impedance levels. Howwee, thfe gives
difficulties for meas”=ements requiring wide bandwidths
because of the RC time constant associated with a
high-megohm resistor and even a few picofarads of cir-
cuit capacitance.
Figure 4 shows a c”==ent s”“=ce
generating a voltage across a parallel RC. The fre-
quency response of this current measurement is limited
by the RC time constant.
Figure 5 shows this response
end the -3 dB p”int “CC”=B at a frequency
Lor* noise and high speed, therefore, a=e contradictory
requirements. TO optimize a current-measuring system,
techniques must be used which obtain high speed “sing
high-impedance devices.
C. Hiah Speed Methods.
1. High epeed can, af c”“=se, be obtained in .
shunt-type meae”=eme”t by “sing a low value for the
shunt resist”=. As pointed ““t above, such a srmll
resistor value int=ad”ces excessive noise into the
meQ8”rement.
2. A second method to achieve bandwidth is to
keep R large, to accept the frequency roll-off
starting at F”, and t” change the frequency eesponae
of the voltage amplifier a8 ehown in Figure 6a. The
combined effects of the RC time c”“sta”c folloved
by this amplifier is shown in Pigure 6b and it is
seen that the frequency response of the c”==ent
measurement has been extended to Pl. The frequency
at which the amplifier gain sta=‘ts to increase must
be exactly equal t” the frequency F” determined by
the RC time c”nstant in order for this approach t”
result in a flat frequency respanse. Therefore,
FO
LOG FREQUENCY
FIGURE 5. The frequency respanse of the shunt method
1s limited by omnipresent ahunt capacitance.
I 0471

MODEL 427 OPERATION
this oethad is ueeful only far application* where
the shunt capacitance C is constant. Aaide from
thin drawback this is * 1eSitimste approach which
is being wed in low-noise, high-speed current-
meesuring applicatians. In addition to current noise
in the *hunt and in the amplifier input stage, B
maJor source of noise in this system *ri*** from
the voltage-noise generator ssaociated with thb in-
put atage (reflected a* current noise in the shunt
resistor) caused by the high-frequency peeking in
the following stages of amplification. More will be
said about this in the discussion on noise behavior.
3. A third method used for speeding up * current
measurement asas guarding techniques to eliminate
the effects of capacit*nces. Unfortunately only
certain type* of capacitance*, such ** cable cap=-
itances, can be conveniently eliminated in this
manner. To eli,r,inate the effect of parasitic c*p*c-
itences associated with the *ource itself became*
very cumbersome and m*y not be feasible in many in-
stsnces. The major *ourc** of noise in this *“*tern
*re identical to those mentioned in the second
system.
4. A fourth circuit configuration combines the
capability of low-noise and high-speed performance
with tolerance for varying input C and eliminate*
need for separate guard by making the ground plane
*n effective guard. This is the current-feedback
technique. This technique gives * typical improve-
ment of 3 over shunt technique*. Again, the major
sources of noise are identical to those mentioned
in the second system.
d. Noise in Current Measurements. Noise forms *
b*aic limitation in *nv hinh-speed current-measurinn
system. The shunt *y&m give; the simplest curren;
measurement but does not give low-noise performance.
A properly designed feedback *y*tem gives superior
noise - bandwidth performance. Noise in these two
systems will be discussed next.
1. Noise Behavior of the Shunt System. High
speed end low noise *r* contradictory requirements
in any current meesurement because *orw capacitance
is always present. The theoretical performance
limftetion of the shunt *yetem c*n be calculated **
FO F> LOG FREWENCY
FIGURE 6. ny tailoring the frequency response of the
amplifier (Pig. 6a) the frequency response
of the shunt method c*n be extended.
The rms thermal noise current (in) generated by *
resistance R is given by
Eq. 4
The equivalent noise bandwidth (.f) of * parallel SC
combination is Af = 1/(4RC) snd the eignal hand-
width (3 dB bandwidth) F, = 1/(2nRC). For practical
purposes peak-to-peak noise is taken 88 5 times the
ml* value. The peak-to-peak noise current can now
be written a*
i
UPP = 2 x 10-9 F, F Eq. 5
In practice, e typical value for shunt cape.cit*nce
is 100 picofarads. With this value the following
rule-of-thumb is obtained. The lowest ratio of
detectable current divided by signal bandwidth using
*hunt-techniques is 2-10-14 ampere/Hertz for B peak-
to-peak signal-to-noise ratio equal to 1. A coroll-
ary far this rule-of-thumb expresses the noise cur-
rent in term* of obtainable risetime (lo-SO% rise-
time tr = 2.2 RC). The lowest product of detectable
current and risetime using shunt technique* is 7 x
lo-l5 ampere seconds. In this derivation it has been
assumed that the voltage amplifier does not contri-
bute noise to the measurement.
2. Noise Behavior of the Feedback System. There
are three *ource* of noise in the feedback system
that have to be looked at closely. The firat two,
input-stage shot noise and current noise from the
mea*urinS resistor, are rather straight-forward. The
third, voltage noise from the input device of the
amplifier, cau*e* *ome peculiar difficulties in the
measurement. Any resistor connected to the input
injects white current noise (Eq. 4). In the circuit
of Figure 7 the only resistor that is connected to
the input is the feedback resistor R. As in the
shunt system, R must be made large for lowest noi*e.
Beceuse this noise is white, the total contribution
can be calculated by equ.,ting Af to the equivalent
noise bandwidth of the system. The second *ource of
noise is the current noise from the amplifier input.
This component is essentially the shot noise asaoci-
ared with the gate leakage current (io) of the input
device. Its rms value equals . . .
FO F, LOG FREQUENCY
FIGURE 6b. Extended frequency response.
0471

OPERATION MODEL 427
T;; = J-zTp-
where e is the electronic charge. The contribution
of this noise generator is also white. N*t only do
these two noise sources generate white current noise,
the noise in a given bandwidth is also independent
of the input capecitence C. The mejor source of
noise in e feedback current meesurement is the noise
contribution aseocisted with the voltage noise of
the input amplifier. The voltage noise ten be rep-
resented by a VOltage noise generator (0,) et the
emplifier input es shown in Figure 8. This wise
generator itself is assumed to be white. However,
its total noise contribution to the current-measuring
system is not white. Inspection of Figure 6 will
reveal that et low frequencies P large em*u*t of feed-
beck ie applied around the voltage noise source {en).
However, the SC combination ettenuetes the high-
frequency components of V,,t so that no feedback is
present et high frequencies. Thus, the noise con-
tribution to the output voltage V,,t from the valt-
age noise source a* is no longer independent of
frequency. The noise is “colored” and increases in
intensity for ell frequencies higher than F,. The
resulting noise spectrum is shown in Figure 9b. The
tote1 system noise is related to the are* under
this curve. Because the logarithm of frequency is
plotted on the horizontal axis, the eree under the
curve et higher frequencies represents e signifi-
cantly larger amount of noise then e similar eree
*t low frequencies. For comparison, Figure 9a show
the frequency response of the current measuring
system. Figure 9e ia identical to Figure 6b. It is
interesting et this point to compare this noise
spectrum with the frequency response of the voltage
amplifier in Figure 4 es shown in Figure 6a. A volt-
age noise eouec.e et the input of the amplifier would
generate a noise spectrum according to the amplifier
frequency response as shown in Figure 6a. The noise
spectrum of such e system, then, is identical to the
noise spectrum of the feedback system as given in
Figure 9h. This illustrates the well-known fact
that signal-to-noise performance of a measurement
cen**t be improved by feedback techniques. At the
high-frequency end the voltege noise is limited by
the frequency FA which is the high-frequency roll-
off point of the operational amplifier. It should
FIGURE 7. Beslc circuit configuration for the feed-
back method.
he noted that even though the useful bandwidth of
the system extends only t* Fl, there era noise com-
ponents of higher frequency present. To obtain
best widebend-noise performance, these high-frequency
noise components have to be removed. This ca* be
achieved by adding a low-pass filter section follow-
ing the feedback input stage. If the band-pass of
thin low-pass filter is made adjuetable this filter
can nerve the dual purpose of removing high-frequency
noise end of limiting the signal bandwidth of the
system.
2-2. THEORY OF OPERATION.
8. Current Feedback Technique. The basic circuit
configuration used in the current-feedback technique
is shown in Figure 7. In this configuration the
current-measuring resistor R is placed in the feedback
loop of e* inverting emplifier with a gain of A*. The
frequency response obtained with this circuit is iden-
tical to thet s+nvn in Figure 6b. F* agein is the
frequency associated with the RC time constant:
F, = SE Eq. 6
The frequency response of the syetem is extended t* a
frequency fl where
F , = AoF, Eq. 7
Note that the frequency rerponse is automatically flat
without heving to match break points. However, the
total bandwidth of the system (Fl) is still limited
by the value of the ahunt capacitance C across the
input. This improved frequency response of the feed-
back technique avoids the use of low values for R
which could generate exceesive current noise.
.
b. Refinements of the Feedback System. A major
difficulty of the feedback system ariees from shunt
capacitence esaociated with the high-megohm resiaeor R
in the feedheck path. If the shunt cepacitence acroes
the resistor is CFr then the bandwidth (FF) of the
system is determined by the time COnstent RCF:
FIGURE 8. The voltage noise associeted with the am-
plifier input device is en important eourc~
of noise in the high-speed feedback syatew
6 0471

MODEL 427 OPERATION
FIGURE 9. The bendwidth of the high-speed feedback
system (Fig. 9a) ten he limited by using
e filter with either e -6 dB/actave or a
-12 dB/octave roll-off. The effect of the
filter on the noise spectrum is showwin
Fig. 9h. Effect of input capacitance on
noise is shown in Fig..9c.
FIGURE 9b. Effect of filter on noise spectrum.
FIGURE 9c. Effect of input capacitance on noise.
0471
FIGURE 10. Frequency compensation.
FF = 1 2nllcp Eq.
A slight modification of the feedback loop can correct
this problem es shown in Figure 10. If the time con-
stant RlCl is made equal to the time constent R.CF,
it CB* be shown that the circuit within the dotted
line behaves exactly es a resistance R. The matching
of time constants in this cese does not become e draw-
beck because the copscitances involved era all constant
and not effected by input impedance.
C. -12 dB/actave Filter.
1. Theory. To obtain optimum widehand noise per-
fomence e filter with e single high-frequency roll-
off (i.e., -6 dB/octave) is not sufficient end -12
dB/octeve is required. The effect of e -6 dB filter
is shown in Figure 9a end h. The filter is used to
limit the system bandwidth to a frequency F2, smaller
then Fl. The effect af this filter on the noise
spectrum is shown in Figure 9b. It ten be seen that
there ace egain high-frequency noise components above
F2, the useable bandwidth of the system. These can
he eliminated by using e filter with e -12 dB/octave
roll-off. The result of such .a filter on noise per-
formance is also shown in Figure 9b.
2. Model 427. The input smplifier 18 followed by
en adjustable low-pass filter having e -12 dB/octeve
roll-off end a valtage gain of 10X. The voltage
gain in the low-pass filter avoids premature over-
loading in the input amplifier which ten be seen es
fallows. The maximum output voltage V,,t is $10
volts. The maximum signal level et the input of the
low-pass filter is, therefore, +l volt. At this
point in the circuit, wide-band noise could still be
present end exceed the l-Volt signal level. The
voltege gain of 10 in the filter allows the total
pre-filter wide-hand noise to exceed the full scale
signal by e factor of 10 (20 dB). The frequency re-
sponse of this filter is edjustahle for variable
“damping” control.
7

OPERATION MODEL 427
2-3. CONNECTIONS,
8. Input. The input receptacle (5202) is e SNC
type which metes with coaxial cables such es Keithley
Models 8201 end 8202. The inner contact is circuit
high. The outer shell is low or chassis ground.
b. Output. Two outp,ut receptacles *re provided
(5102 on the front, 5103 on the reer panel). These
era BNC type* where the inner contact is output high
end the outer shell is chassis ground.
2-5. OPERATING CONSIDERATIONS.
8. Gain. The gein of the Model 427 is defined in
terms of volts per *“pare. Since the output level is
10 volts for e full scale input, the gain could also
be expressed e* sensitivity in emperes referred to the
input BS in Table 2-1. Vout = - (Iin x GAIN) Eq. 9
TABLE 2-l.
Gain or Sensitivity Referred to the Input
C. Power Input. The power receptacle (P305) on the
rear panel is a 3-prong connector which metes with
Keithley pare number CO-6 line cord.
2-4. CONTROLS.
a. Front Panel.
1. Power Switch “PUSH ON” (5302). This switch
controls the line power to the instrument. ‘The
switch is a special pushbutton type with “Power On”
indicated by a self-contsined pilot lamp.
2. GAIN (VOLTS PER AMPERE) (S201). This switch
sets the overall gain in eight positions from 104
to loll. A “ZERO CHECK” position permits adjust-
ment of zero offsets.
3. RISE TIME Switch (5101). This switch sets
the lo-90% rise time in 10 positions from .Ol to
300 milliseconds(for the filter section only).
4. SUPPRESSION (MAX) Switch (S303). This switch
sets the maximum current suppression in eight pasi-
tions from lo-10 to 10-3 A. When the switch is set
to “OFF” the current suppression circuit is disabled.
5. SUPPRESSION (FINE) control (~333). This con-
trol permits adjustment of suppression with 0.1%
resolution.
6. SUPPRESSION (POLARITY) Switch (S304). This
switch set* the polarity of the current suppression
(referred to the input).
7. ZERO ADJUST Control (R235). This control per-
mits adjustment of zero offset through the u*e of the
OVERLOAD indicator.
b. Rear Panel.
1. Line Voltage Switch (S301). Sets instrument
for either 117 or 234 V operation.
Fuse RequirP.ment* 3AG, Sla-Slo
117V: 1/4A Keithley No. F”17
234V: l/SA Keithley No. F”-20
GAIN Feedback
Setting Resistor
Full Scale Full Scale
sensitivity Output
(Amperes) (Volts)
103 1
104 x 10-3
1 x 10-4
105 1 x 10-5
106 I x 10-6
107 1 x 10‘7
108 1 x 10-g
109 1
1010 x 10-9
1 x 10-10
10
10
10
10
10
10
10
10
b. Rise Time. The rise time for each gain setting
is listed in the specifications a* “FAST RISE TIME”.
These rise times are obtained when the RISE TIME
switch is set to the positions indicated in Table 2-2.
TABLE 2-2.
Switch Settings for “FAST RISE TIME”
GAIN Rise
Setti”g Time
104
105 15 ps
106 15 us
107 15 I*8
40 us
RISE TIME
Setti”g*
.Ol “S
.Ol ms
.Ol “S
.03 “9
DYlWl”iC
Range
2000
2000
2000
2000
1 10; 60 ps .03 ms 800
1oy 220
1010 &Is .l “S 400
loll 400 “Sps .3 ms 200 1.5 1 ms 100
J
c. Suppressian. Current suppression is provided in
the Model 427 for suppression of input currents up to
10e3 amperes. By suppressing background currents, smell
variations in e larger signal can be observed. Currents
of either polarity can be suppressed. To suppress an
input current the SUPPRESSION should be *et to supply
e current of apposite polarity. The FINE control permits
adjustment up to 1.5 times the MAX setting.
d. Overloads, The overload sensing circuit detects
en overload et two places in the circuit: before end
after the “RISE TIME” filter circuit. The OVERmAD
lamp (DS302) will indicate whenever the voltsge sensed
ia greater then full scale regardless of the RISE TIME
setting or the frequency.
a. Zero Adjust. The ZERO CHECK po*ition grounds the
the input of the instrument and co”“ert* the cuerent
amplifier to a high-gain voltage amplifier. The ampli-
fied offset voltage will turn on the OVERLOAD indicator
whenever the input voltage offset exceeds 5100 t0l.
Therefore the ZERO control should be adjusted so that
the OVERLOAD indicator is off when in ZERO CHECK mode,
yielding the specified input voltage drop.
8

MODEL 427 APPLIChTIONS
SECTION 3. APPLICATIONS
3-1. CURRENT~MEASURING SYSTEM. The typical current
meaeuring system consists of a current source, a cur-
rent amplifier, and a monitoring device, The current
source could include an ion chamber, photomultiplier,
or other high-impedance device, The current amplifier
such as the Model 427 provides sufficient gain to drive
a monitoring device such as a chart recorder or other
readout. The Model 427 in this case provides an out-
put voltage which is calibrated in volts per ampere
a.3 in equation 10.
Iin = - (V,,t I GAIN)
Eq. 10
Example : GAIN = 106 voltslampe~e
” O”t = +500 In”
The input current Iin would be:
lin = - (5x10-~vo1ts/106vo1ts per ampere)
Iin = - 5 x 10-7 amperes
3-2. NOISE BANDWIDTH CONSIDERATIONS. Table 3-l illus-
trates the trade-off between fast rise time and dynamic
range. For this application dynamic range is defined
a.s the ratio of maximum peak-to-peak current to peak-
to-peak current noise. Peak-to-peak current noise is
taken as 5-times the rms current noise. The maximum
peak-to-peak current is 2-times the maximum full scale
current. NOTE
When using current suppression the current-suppres-
sion resistor should be considered as an additional
current-noise generator. The values given in Table
3-l do not include the contribution of the suppres-
sion resistor. Therefore the selected suppression
resistor Rs, should be as large as possible to min-
imize the contribution to current noise.
3-3. NOISE-IMPROVEMENT CONTOURS. The sensitivity
and speed of the Model 427 (for either d-c or a-c
measurements) can be compared to the best perfarm-
ante obtainable with the shunt method of measuring
current. The best “noise-risetime” product that
can be achieved for d-c measurements (with 100 pF
shunt capacitance) in a shunt system is 7 x lo-15
ampere-seconds. However the feedback system achieves
2 x lo-l5 ampere-seconds (also with 100 pF shunt
capacitance). When used in a-c narrowband systems
(lock-in, etc.) the-degree of improvement is a func-
tion of shunt capacitance and operating frequency.
The achieveable imprownene over the shunt method
can be plotted in a graph similar to a set of noise
contours. Figure 11 shows the measured impravemene
(negative dB) that can be obtained with the Model
427 at a given frequency and shunt capacitance when
compared to an ideal (noiseless) amplifier Fn a shunt
system.
,
I FREOUENCYIHI1
FIGURE 11. Plot of noise-improvement contours.
TABLE 3-l.
RMS’Noise Current (Typical)1 as a Function of Gain and Rise-Time Setting
,vL..I,.,“,.,
PULL SCALE
GAIN CURRENT RISE TIME SETTING
VIA AMPERES 300 100 30 10 3 1 .3 .l .03 .Ol
104 10-3 * * * * * * 1x10-8
105 10-4 1.2x10-8 4x10-8 1x10-7
* * * * * *
106 1x10-9 1.2x10-9 4x10-9 1x10-B
10-5 * * * * *
107 m-6 : * * * * lx& 1x10-10 1.2x10-10 4x10-10 1x10-9
108 10-7 1.5x10-11 2x10-11 1x10-10 x
* * *
109 MO-12 2x10-l2 5x10-12 1x10-11
104’ * 4x10-11 x
1010 10-9
lo-1o * 1x1:-14 *x1:-14 2do-13 2d0-13 5do-13 2~0-12 5x10-12 x x
x x
1011 5~0-14 2do-13 ~0-13 2do-12 x
2x10-15 4x10-15 1x10-14 4x10-14 1x10-13 4x10-13 x x x x
1 With up to 100 pP input shunt capacitance. Noise increases aa input shunt capacitance increases.
KEY :
x = Filter Bandwidth is greater than current-amplifier bandwidth.
* = larger Rise Times are useful for increased filtering of the signal arid noise inherent in the source.
They do not further improve the instrument noise contribution except when the input shunt capacitance
exceeds 100 DF.
0471 9

ACCESSORIES
SECTION 4. ACCESSORIES
MODEL 427
4-l. GENERAL. The following Keithley accesaaries can 4-2. OPERATING INSTRUCTIONS. A separate Instruction
be used with the Model 427 eo provide additional con- Manual is supplied with each accessory giving complete
venience and versatility. operating information.
Model 1007 Rack Mounring hit
Description:
The Model 1007 is a dual rack mounting kit with over-
all dimensions 3-l/2 in. (64 mm) high and 19 in. (483
mm) wide. The hardware included in this kit consists
of two Angle Brackets, one Mounting Clamp, and extra
mounting BCreWS.
Application:
The Model 1007 co""erts any half-rack, style "M"
instrument from bench mounting to rack mounting in
a standard 19-inch rack. The kit may also be used
for rack mounting 19-inch full rack width insfru-
ltE"tS.
The Model 1007 Rack Mounting Kit can be used to m"u"t
instruments of 11 inch or 14 inch depth. The user
should decide the position af the i"~tr"me"ts to be
rack mounted. The Assembly Inaeructions refer co
instruments positioned as shown and identified as
instrument “A” and "B".
Parts List:
Item
NO. DWCl+ltiO”
22 Angle Bracket
23 Screw, 16-32 x 5/8,
Phillips Pa" Hd
24 Mounting Clamp
25 Screw, %6-32 x 1,
Phillips Pa" Hd
26 Kep Nut 116-32
27 Screw, 116-32 X l/2,
Phillips Pa" Hd
28 Screw, 116-32 x 718,
Phillips Pa" Hd
VY Keithley
Keq'd Part No.
2 27410B
6 -_
1 247988
1 __
3 _-
2 __
1 __
10 0877

MODEL 427 ACCESSORIES
Model 1007 Dual Rack MountinS Kit
Assembly Instructions:
1. Before assembling the rack kit, determine the
pasition of each instrwnent. Since the inserumenfs
can be mounted in either location, their position
should be determined by the user’s meas”rement. The
following instructions refer to instruments “n” and
UB” positio”ed as shorvn. For mounfinS 19-inch full
rack Width instruments, disregard steps 2 through 5.
2. Once the position of each instrument has been
determined, ebe “side dress” panels on both sides of
each instrument should be removed. Renxwal is
accomplished by looseninS the screwy (Item 8, oriS-
inal hardware) in two places. Slide the “side dress”
panels co the rear of the instrument to remove.
3. The mountinS clamp is installed on instrument
“A” using the oriSina1 hardware (Item 8). With the
screws removed, insert the “mounting clamp” behind
the “corner bracket” (Item 7) and replace the screws
to hold the mounting clamp in place.
4. Tighten the screwy (Item 8) on instrument “B”.
Insert the “mounting clamp” behind the “corner
bracket” (Item 7) an instrument “8” a8 shown.
5. When mounting instruments having the same depth,
a screw (Item 25) and kep nut (Item 26) are required
to secure the two instruments together. When ,,,oune-
ing instruments of different depth, da not use kep
nut (Item 26) but substitute shorter screw (Item 28).
6. Attach an “anSle bracket” (Item 22) on each
instrument using hardware (Item 23) in place of the
original hardware (Item 8). For 14 in. long instru-
ments use 116-32 x 518 Phillips screw (Item 23) with
116-32 kep “uf (Item 26).
7. The bottom cover feet and tilt bail assemblies
may be removed if necessary.
8. The original hardware, side dress panels, feet
and tilt bail assemblies should be retained for fut-
ure conversion back to bench mountine.
0777 11

CIRCUIT DESCRIPTION MODEL 427
SECTION 6. CIRCUIT DESCRIPTION
5-1. GENERAL.
The Made1 427 is composed of a feed- a. +15 ” Reguletor. AC power ia tapped from one
beck amplifier, e X10 gain filter section, suppression secondary of transformer T301. The ec ia rectified by
.“d power supply circuits ee show” in Figure 12. The P full-weve bridge rectifier (D301). Trensistor Q301
feedback emplifier is located an the “Amplifier Beerd”, is the series pees reguletor. Integrated circuit
QA
PC-289. The filter circuitry is located on the “Pilter 301 is . self-conteined reference end regulating cir-
Board”, PC-291, PC-292. The power supply circuitry is alit. Potentiometer R304 is a” internal voltage ed-
located on the “Mother Board”, PC-290. justment. Resistor R307 serves as a current limit de-
vice.
5-2. FEEDBACK AMPLIFIER (PC-289). The feedback empli-
fier is composed of e high-gain amplifier connected ee
e feedback emmeter. The feedbeck resietors R22o through
R227 are set by the GAIN Switch (SZOl). The high-gain
emplffier is composed of a dual FET input stage (Q20IA
end B), e differential amplifier (QAZOl), end en output
stage (9203 end Q204). The feedback is connected from
resistor R201 to the cutput stage et Q203 end Q204.
Potentiometers R232, R233, and R234 ere internal fre-
quency compensetion controls for leg, lOlo, end 1011
gains respectively. Potentiometer R235 is the ZBRO
AD., cantrol. The full scale output of the feedback
emplifier is 1 volt.
5-3. PILTBR (PC-292). The filter circuit ie composed
of e high-gain amplifier connected aa e 12 dB/octave
low pass filter es sham in Figure 13. The amplifier
consists of intel(rated circuit QAlOl end ~“tp”t stege
(QlOl end Q102).- Full scale output ie 10 ;olts. -
The gain is established et X10 by resistors RllO end
R112 + R113. Potentilnwtel: RlOS is en internal zero
adjustment. Potentiometer R113 is en interns1 gain
adjustment.
b. -15 ” Regulator. AC power is tapped from one
secondary of trenaformer T301. The ec is rectified by
P full-wave bridge rectifier (D302). Trensietee Q302
is the series pees reguletor. Integrated circuit QA-
302 is e self-contained reference end regulating cir-
cuit. Potentiometer R309 is en internal voltage ad-
,ustment. Resistor R307 mrves as a current limit de-
vice.
5-5. CURRKNT SUPPRESSION CIRCUITRY (PC-290). The
suppression is applied et the input es show” in Figure
15. The N&X AMPERES Switch (5303) sets the current
suppression in decade steps from lo-3 te lo-lo *Slp.3=.3*
(Resistors R325 through R332). Potentiometer R333 is
the FINE Control which provides adjustment frnn 0 to
1.5 times the WAX setting. Switch S304 eete the pol-
arity (either + end - 15 volt eource). Current sup-
presaionis e function of V,,&S,
--
where “cS = Voltage et the wiper of R333.
IQS = Series Resistor (R325 through R332).
Exemple: If MAX AMPEP.Rs = 10-b
5-4. POWER S"PPLY (E-290). The pewer supply provides
+15 V dc et up to 70 mA for the amplifier circuits.
The regulator circuits ere composed of identicel cam-
ponents end are connected ee shown in Figure 14.
end if “cS = cl5 ”
the* Its = +15v = +1.5 x 10-G amperes
107n
GAIN RISE TIME
III
r-5
I I
FILTER
2!5
SUPPRESSION
12
FIGURE 12. Block die‘rr of l hi‘h-‘peed current l mplifier 0471

Cl01
I,
RIOI r--------1
0
RI16
FIGURE 16. component Layout - PC-291.
0471 13

COMI
14
1
FIGURE 17. Component Layout - PC-290.

MODEL 427
?
COMPONENTLAYOUTS
FTGURE 18.
component Layout - PC-289. FIGURE 19.
component Layout - PC-292.
0471 15

COMFUNENTL4YOoTS MODEL 427
PC-290
FIGURE 20. Chaaais - Top View.
16 0471