Moog Norlin 204D User manual

TABLE OF CONTENTS
SPECIFICATIONS .
1 INTRODUCTION .
1.1 GENERAL.
1.2 PRECAUTIONARY MEASURES .
2 CIRCUIT DESCRIPTION .
2.1 GENERAL.
2.2 GENERAL CIRCUIT DESCRIPTION.
2.3 "D" OSCILLATOR PRINTED CIRCUIT BOARD.
2.4 MODULATION MIX AMPLIFIER.
2.5 NOISE GENERATOR.
2.6 HEADPHONE AMPLIFIER .
2.7 POWER SUPPLY.
2.8 VOLTAGE CONTROLLED FILTER .
2.9 VOLTAGE CONTROLLED AMPLIFIER.
2.10 CONTOUR GENERATORS .
2.11 A-440 REFERENCE OSCILLATOR .
2.12 EXTERNAL PREAMPLIFIER AND OVERLOAD LAMP DRIVER.
2.13 OLD MINIMOOG OSCILLATOR BOARD .
2.14 -5 VOLT REFERENCE SUPPLY .
2.15 OCTAVE SWITCHES .
2.16 OSCILLATOR ONE .
2.17 OSCILLATOR TWO .
2.18 OSCILLATOR THREE.
2.19 KEYBOARD CIRCUIT..
3 TROUBLESHOOTING .
3 1 OVERALL QUICK REFERENCE TROUBLESHOOTING .
3.2 O SCILLATOR PRINTED CIRCUIT BOARD NO. 1 TROUBLESHOOTING
(SERIAL NUMBERS 10176 AND ABOVE).
4 DISASSEMBLY PROCEDURES.
41 DISASSEMBLY .
5 ADJUSTMENT AND TUNING .
5.1 PRINTED CIRCUIT BOARD REPLACEMENT ADJUSTMENTS.
5.2 VOLTAGE ADJUSTMENTS.
5.3 TUNING OSCILLATORS (SERIAL NUMBER 10175 AND ABOVE) . .
5.4 TUNING THE OSCILLATOR (SERIAL NUMBERS 8ELOW 10175) . . .
5.5 DETAILED ADJUSTMENT PROCEDURES.
5.6 POWER SUPPLY ADJUSTMENT .
5.7 A-440 ADJUSTMENT.
5.8 SIGNAL FLOW .
5.9 MAIN OUTPUT .
5.10 PHONE OUTPUT.
5.11 LOW MAIN OUTPUT .
5.12 EMPHASIS CALIBRATION.
5.13 FILTER CUTOFF ADJUSTMENT.
5.14 FILTER SCALE.
5.15 KEYBOARD 1/3.
5.16 EXTERNAL FILTER CONTROL .
5.17 AMOUNT OF CONTOUR . . .
5.18 FILTER CONTROL.
5 19 AMOUNT OF MODULATION .
5.20 FILTER CONTOUR.
5.21 DOUBLE TRIGGERING .
5.22 LOUDNESS CONTOUR .
5.23 EXTERNAL LOUDNESS CONTROL UNIT.
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TABLE OF CONTENTS (Continued)
SECTION PAGE NO.
5.24 NOISE . 5 10
5.25 CONTOUR GENERATOR BALANCE. 5-10
5.26 EXTERNAL SIGNAL INPUT . 5-10
5.27 AUDIO NOISE GENERATOR . 5-10
5.28 GLIDE . 5-10
5.29 KEYBOARD PITCH CONTACTS . 5-11
5.30 OSCILLATOR ADJUSTMENTS FOR BOARD. 5-11
5.31 OSCILLATOR RANGE . 5-11
5.32 OSCILLATOR 1 TUNING. 5-11
5.33 OSCILLATOR 2 AND 3 TRACKING. 5-11
5 3 4 OCTAVE ADJUST . 5-11
5.35 RANGE OF TUNE. PITCH AND FREQUENCY ADJUST CONTROLS. 5-11
5.36 OSCILLATOR 3 WIDE RANGE. 5-12/6-0
5.37 MODULATION OF OSCILLATOR. 5-12/6-0
5.38 EXTERNAL CONTROL OF OSCILLATOR. 5-1 2/6-0
6 KEYBOARD MAINTENANCE . 5-12/6-0
6.1 CLEANING . 5-12/6-0
6.2 KEYBOARD ADJUSTING PROCEDURE . 6-1
7 REPLACEMENT PARTS LIST . 7-0
7.1 ORDERING . 7-0
8 MODIFICATIONS . 8-1
8.1 SERVICE BULLETINS. 8 1
8.2 OSCILLATOR BOARD ASSEMBLY . 8 1
8.3 CONTOUR GENERATOR ASSEMBLY . 8-1
8.4 POWER SUPPLY ASSEMBLY . 8-1
8.5 FILTER ASSEM8LY . 8-1
8.6 KEYBOARD CIRCUIT PRINTED CIRCUIT BOARD NO. 2 . 8 1
8.7 MINIMOOG OSCILLATOR TUNING. 8-2
8.8 OSCILLATOR BOARD NO 1. POWER SUPPLY CONNECTION AND
OCTAVE BUFFER. 8-2
9 SCHEMATIC DIAGRAMS AND PRINTED CIRCUIT BOARDS . 8-8/9-0
LIST OF ILLUSTRATIONS
2-1 Simplified Block Diagram . 2-11
2-2 Subassembly Location Diagram . 2-12
2-3 Printed Circuit Board Socket Location. 2-12
2-4 Left Hand Controller . 3-1
5-1 Power Supply Adjustment Location Diagram . 5-2
5-2 Rear Panel Tuning Control Location Diagram (Serial Numbers 10175
And Above). 5-2
5-3 Filter Adjustment Location Diagram .-. 5-2
5-4 Front Panel Control Settings (Serial Numbers 10175 And Below) . 5-2
5- 5 Front And Rear Panel Location (Serial Numbers 10175 And Below) .... 5-2
6- 1 Use Of Keyboard Adjusting Tool. 6 1
7- 1 Front Panel Control Diagram . 7-1
8- 1 Printed Circuit Board 2 Modification . 8-3
8-2 Rear Panel Modification Diagram. 8-4
8-3 Front Panel Controls Diagram. 8-5
8-4 Rear Panel Adjustment Location Diagram . 8-5
8-5 Connector Modification Diagram. 8-7
8-6 Octave Buffer Schematic. 8-8/9-0

POLYPEDAL MODEL 285A

Specifications
SOUND SOURCES
NO. OF SOUND SOURCES: 5 (3 Oscillators, 1
Noise Source, 1 External Input/Microphone Pre-
OSCILLATOR FREQUENCY: 0.1 to 20kHz
(cycles/second) in six overlapping ranges.
SHORT TERM OSCILLATOR STABILITY:
Better than 0.25%.
OSCILLATOR WAVEFORM OUTPUTS: Tri¬
angular, Sawtooth, Triangular-Sawtooth Mix (Oscil¬
lators 1 and 2 only). Reverse Sawtooth (Oscillator
2 only), 3 widths of Rectangular.
NOISE SOURCE OUTPUTS: White or Pink
random waveforms.
PREAMPLIFIER INPUT: 10 millivolts mini¬
mum: 2 volts maximum.
PREAMP INPUT IMPEDANCE: 100K ohms or
greater.
FILTER
FILTER CHARACTERISTIC: Wide-range low-
pass filter with variable-height resonant peak at
cut-off frequency, and 24dB/octave cutoff slope.
RANGE OF CUTOFF FREQUENCY: Con¬
tinuously variable from 40Hz to 20kHz (9 octaves).
VOLTAGE CONTROLLED AMPLIFIERS
NUMBER OF AMPLIFIERS: 2 (one controUed
only by its Contour Generator; the other con¬
trolled by optional external controller).
DYNAMIC RANGE OF EACH AMPLIFIER:
80dB.
CONTOUR GENERATORS
NUMBER OF CONTOUR GENERATORS: 2
(one controlling filter through an attenuator; the
other controlling the first voltage Controlled
Amplifier).
RANGE OF ATTACK TLME: 10 milliseconds
to 10 seconds.
RANGE OF DECAY TIME: 10 milliseconds to
10 seconds.
RANGE OF SUSTAIN LEVEL: 0 to 100% of
contour peak.
WIDTH OF SWEEP OF FILTER BY ITS CON¬
TOUR GENERATOR: Continuously variable from
0 to 4 octaves.
AUDIO SIGNAL OUTPUTS
HIGH LEVEL OUTPUT: 0.5 volts typical,
with 3K ohms nominal output impedance.
LOW LEVEL OUTPUT: 15 millivolts typical,
with IK ohm output impedance.
HEADPHONE OUTPUT: 0.3 volts maximum,
into standard 8-ohm stereo headphones.
CONTROLLERS
KEYBOARD FUNCTION: Permanently con¬
nected to (a) control Oscillators 1 and 2, and (b)
trigger Contour Generators. Keyboard may be
switched to control Oscillator 3 and Filter.
DESCRIPTION OF KEYBOARD: Standard 44
key (3-1/2-octave) organ keyboard. Only lowest
key depressed has effect in controlling Oscillators
and Filter. Contour Generators are activated when¬
ever a single key is depressed.
RATE OF KEYBOARD GLIDE: Continuously
variable from 1 millisecond to 1 second/octave.
PITCH BENDING RANGE: 5 semitones mini-
MODULATION INJECTION RANGE: 0 to
1-1/4 octaves.
CONTROL AND POWER CONNECTIONS
EXTERNAL PITCH CONTROL INPUT CHAR¬
ACTERISTIC: 1 volt change produces 1 octave
frequency change, + 2 percent.
EXTERNAL FILTER CONTROL INPUT: 1
volt change produces 1 octave change in cutoff
frequency, + 5 percent.
EXTERNAL AMPLIFIER CONTROL INPUT:
Linear control voltage/gain relationship. Gain range
spanned by 0-4 volts.
EXTERNAL TRIGGER INPUT: Switch-closing
activates both Contour Generators.
AUXILIARY DC POWER SOCKET: +10 volts
and -10 volts at 50 mill am peres.
DIMENSIONS AND WEIGHT
OVERALL SIZE (with Front Panel down):
28-5/8 inches (72cm) wide, 17-1/8 inches (41cm)
deep, 5-3/4 inches (14cm) high.
NET WEIGHT: 28 pounds (12.7kg).
SHIPPING WEIGHT: 45 pounds (20.5kg).
POWER REQUIREMENTS
100-135 and 200-270 volte, 50-60Hz, 40 watts
maximum. Specifications subject to change.

major differences in the two boards is that common
summing is used on the new board while individual
summing is used on the old board.
All three oscillators have individual octave
switching signals applied to three adjustable poten¬
tiometers providing precision octave switching of the
individual oscillators.
2.2.2 MODULATION MIX AMPLIFIER
A modulation mix amplifier selects the output
of oscillator 3 or noise, or both, sums and routes
them to the amount control in the left hand con¬
troller.
2.2.3 NOISE GENERATOR
The Minimoog contains a noise generator using
a transistor generating white noise in the range of
-60dB which is amplified to produce white, pink or
red noise, selected by the noise selector switch.
White or pink noise is used for audio and pink or red
for modulation.
2.2.4 HEAD PHONES
An amplifier is provided to supply sufficient
current to drive standard stereo headphones.
2.2.5 POWER SUPPLY
A dual output power supply with the rectifier
on Board 3, the filter on Board 5 and the transformer
on the chassis, provides + 10 volt regulated power.
The input is either 115 or 230 volt as selected through
a switch on the top of the panel.
2.2.6 KEYBOARD CIRCUIT
The keyboard circuit consists of a temperature
compensated current source, the actual keyboard and
a track and hold amplifier. The keyboard voltage is
routed to a track and hold amplifier where, with no
keys depressed, the circuit is in a hold mode. With a
key depressed, the pitch voltage occurs before the
trigger and the trigger goes off before the pitch voltage
is released.
2.2.7 A-440 REFERENCE OSCILLATOR
The A-440 reference oscillator is based on a
standard We in Bridge oscillator design. The bridge
has a phase shift of zero degrees at approximately
440Hz, supplying a positive feedback and causing
the circuit to oscillate. The frequency of the oscil¬
lation is set by a resistor. Three capacitors and a
resistor limit the signal amplitude to a transistor
collector resulting in a clean sine wave output.
2.2.8 EXTERNAL AMPLIFIER
The external preamplifier is fed to the EX¬
TERNAL INPUT VOLUME control where the out¬
put is fed through a 200 gain amplifier and in turn
routed to the EXTERNAL INPUT ON OFF switch.
This switch allows an external source to be summed
with other signal sources.
2.2.9 OVERLOAD LAMP DRIVER
The output from the 200 gain amplifier is also
fed to the overload lamp driver circuit. The amplitude
envelope of the signal is detected and fed to a voltage
divider. When the AC input voltage reaches ap¬
proximately 1.2 volts, the OVERLOAD lamp illumi¬
nates and is held for a suitable period.
2.2.10 VOLTAGE CONTROLLED FILTER
“Audio” signals are summed and a differential
signal current is generated and applied to a filter
ladder. Cutoff frequency of the filter is controlled
by the standing current through the ladder. The
higher the standing current, the higher the cutoff
frequency. A differential signal is created and fed to
a gain amplifier which converts the differential input
to a single ended output in tum fed to a transistor
in the voltage control amplifier. The FILTER EM¬
PHASIS control is also connected to the gain ampli¬
fier providing positive feedback at the cutoff fre¬
quency. As the amount of emphasis is increased, the
Q increases and the filter starts a sine wave oscillation
at the cutoff frequency.
The cutoff frequency control current originates
in an exponential voltage to a current converter.
Control voltages from the external input, keyboard
2-2

switches, modulation and the contour generator are
summed and fed to the exponential convertor. A
resistor sets the keyboard scale at one volt per octave
and in turn the front panel filter CUTOFF FRE¬
QUENCY control is calibrated in a similar manner.
Z.2.11 VOLTAGE CONTROLLED AMPLIFIER
The input signal from the voltage controlled
filter is AC coupled to a differential transconductance
multiplier. The gain of the transistor is current
controlled; the greater the current, the higher the
gain.
2.2.12 CONTOUR GENERATORS
Separate contour generators are provided for
both filter and loudness contours. Each contour
generator has its own controls consisting of ATTACK,
DECAY and SUSTAIN’. Contour generators are
triggered by an internal voltage trigger from the
keyboard and/or the rear panel S-TRIG jack. The
amount of contour control adjusts the sweep of the
voltage controlled lowpass filter.
2.3 "D" OSCILLATOR PRINTED CIRCUIT BOARD
(Refer to Figures 9-1 and 9-2)
The new “D” oscillator printed circuit board
is a plug-in replacement card for earlier oscillator
boards. This board contains three precision voltage
controlled oscillators with a variety of control inputs
and a number of waveform outputs for each of the
three oscillators.
The schematic diagram shows not only the
oscillator board itself, but several of the associated
control circuits and output selection circuits for
circuit clarity. A variety of input control signals
namely, PITCH WHEEL, TUNE, MOD WHEEL and
OSCILLATOR (three oscillators) are summed via the
"primary oscillator summer" amplifier IC1A. This
common summing (as opposed to individual summing
sections for these control inputs in the old Minimoog
oscillator printed circuit board) guarantees that when
the oscillators track the keyboard input, they also
track these other common drive signals. The output
of the primary oscillator summer, IC1A, is scaled at
this point to provide 1.00 volt per octave by adjusting
oscillator scale adjust potentiometers R16, R55 and
R91 of the three oscillator sections. These three scale
potentiometers are adjusted so that the three oscil¬
lators on each board track the keyboard drive signal
exactly. Final adjustments of these scale potentio¬
meters must be made for an assembled unit, since
the keyboard scale factor for different frames will
vary slightly.
All three oscillators have individual octave
switching signals applied to the octave adjust po¬
tentiometers R18. R57 and R92. These octave adjust
inputs provide precision octave switching of the
individual oscillators, and as the scale adjusts, must
be adjusted to their final value in the assembled
Oscillator 2 has an additional input labeled
OSCILLATOR-2 FREQUENCY which provides for
tuning + a musical fifth. Oscillator 3 has two addi¬
tional inputs which are both driven by the OSCIL¬
LATOR-3 FREQUENCY CONTROL providing + a
musical fifth control when the OSCILLATOR-3
CONTROL is on (namely, when the keyboard con¬
trol voltage is applied to oscillator 3 by the switch¬
ing FET Q2) or providing + 3 octaves of control
when the OSCILLATOR-3 CONTROL is off. When
this control is off, both feed resistors R96 and R97
are connected to the OSCILLATOR-3 FREQUENCY
front panel potentiometer.
The various frequency control inputs for the
three oscillators are summed together in amplifiers
IC1B, 1C3A and IC3B. Resistors R22, R62 and R100
provide for bias current temperature drift compensa¬
tions. It should be noted at this point that all three
oscillators are identical from this point on, except
for the addition of a reverse sawtooth circuit asso¬
ciated with Q20 in oscillator 3. For this reason we
will discuss the detailed operation of only oscillator 3.
The composite frequency control signal appear¬
ing at the output of IC3B is applied to the precision
voltage divider comprised of R101 and R102 which
divides the composite frequency control signal down
to 21.0mV. octave. This divided down frequency
control signal is applied to the base of the expo¬
nentiating transistor Q17 which is contained along
with its companion transistor Q18 in a tempera¬
ture stabilized package. IC15 (Fairchild 726). This
2-3

integrated circuit is maintained at a constant temper¬
ature of approximately 78.4°C by circuitry internal
to the integrated circuit. The actual temperature of
the chip is set by the resistor R105. Transistor Q18 is
a Vjj C compensation transistor whose collector current
is maintained constant by a combination of R108,
R106 and IC16. The plus input of IC16 is maintained
at ground. The output of IC16 sinks current through
the emitter of Q18 such that the voltage appearing
on pin 2 of IC16 is also ground. If we assume that
there is no bias current flowing into the negative
input of IC16 (valid since this integrated circuit is a
FET input operational amplifier) then we see that
5.0 microamperes of current is flowing into the
collector of transistor Q18. regardless of the current
drawn through the emitter of transistor Q17. This
circuit configuration provides a precision tempera¬
ture compensated base emitter reference subtraction
voltage for the exponentiating transistor Q17. The
current flowing into the collector of Q17 is expo¬
nentially related to the voltage applied to the base on
pin 1. This current is applied to the oscillator inte¬
grating capacitor C29 via the “high end comp”
potentiometer R109. The reference side of the inte¬
grating capacitor C29 is tied to a 5.0VDC reference
rail derived from the +10VDC rail via the resistor
divider network R103 and R104, and the voltage
follower IC14A whose output is bypassed to ground
via C28. The FET Q14 shunts the integration capaci¬
tor C29 to +5 whenever the voltage appearing at the
input of the comparator amplifier IC17 reaches
ground. The voltage appearing on the lower side of
the integrating capacitor C29 is a ramp headed
toward 0.0VDC. The voltage appearing on the inte¬
grating capacitor plus the voltage drop across the
“high end comp” potentiometer R109 is buffered by
the voltage follower comprised of IC17 and emitter
follower Q15, R114. R115 and R116. The buffered
negative ramp appearing on the emitter of Q15 is
applied to the comparator IC18B which switches
the shunt FET Q14 on when the ramp voltage
reaches zero via coupling diode CR5 and capacitor
C31. When the shunt transistor Q14 turns on, the
capacitor voltage is returned to +5VDC which in
turn causes the shunt FET Q14 to turn off via the
buffer amplifier IC17, the comparator 1C18B and
pull up resistor R110. Then the negative going ramp
integration begins again. As the integration current
supplied via R109 is increased, a residual voltage up
and above the linear ramp signal appearing across
R109 shows up at the voltage follower IC17 and
thereby at the voltage comparator IC18B. This
residual voltage means that the actual ramp value
appearing on the capacitor required to trigger the
comparator IC18B is reduced at higher frequencies.
It is posable to set R109 so that the effect of non-
linearities due to a finite reset time of the capacitor
C29 are compensated for by this additional residual
voltage. Resistor R112 and capacitor C34 provide for
a small amount of positive regeneration (Schmitt
trigger action) to insure clean switching during saw¬
tooth reset. The sawtooth voltage appearing at the
emitter of Q15 is applied to the bias network com¬
prised of R114, R115 and R116. The values of this
network are chosen so that the voltage appearing
on the sawtooth output ( pin 13B) is precisely
+1.75VDC to -1.75VDC. Resistor R137 is a selected
resistor whose value is chosen to achieve accurate
symmetry in this output waveform. This symmetry
is important to achieve an accurate 50 percent duty
cycle of the rectangular waveform appearing on
pin 15B.
While the sawtooth appearing on the emitter of
Q15 goes precisely from +5VDC to 0.0VDC, a
voltage equal to V^ e above this is available at the
base of Q15. This base voltage is also applied to the
triangular wave converter transistor Q16. Triangular
conversion circuit comprised of Q16, R118 and R120
acts as a precision inverter as the sawtooth waveform
travels between OVDC and +2.5VDC. From 2.5VDC
to 5.0VDC transistor Q16 is in saturation with
essentially zero collector to emitter voltage drop.
In this condition the emitter of Q16 follows very
closely the emitter of Q15 which, of course, is the
sawtooth waveform. Since there is negligible col-
lector-to-emitter voltage drop in this saturated condi¬
tion, the collector of Q16 then follows accurately
the sawtooth waveform between 2.5VDC and 5VDC.
Therefore, during half of the cycle the triangular
conversion circuit acts as a precision inverter and
during the second half of the cycle it acts as a pre¬
cision voltage follower, resulting in a precision
triangular waveform output. This precision triangular
waveform appearing at the collector of Q16 is then
applied to the buffer with gain comprised of IC14B,
R121 and R122 via the de-glitching RC network
comprised of R119 and C35. This suppresses the
fast transient which occurs in the triangular wave
output which occurs during the sawtooth waveform
2-4

reset. The output of the follower IC14B is a trian¬
gular waveform which goes from +1.75VDC to
-1.75VDC.
The sawtooth waveform appearing on the out¬
put pin 13B is applied to the rectangular wave shap¬
ing circuit comprised of 1C18A, R117, R125 through
R127, and R131 and R132. A control voltage which
ranges between OVDC and -2.5VDC is applied to the
negative input of IC18A via the biased divider net¬
work R126 and R127. The sawtooth waveform is
applied to the positive input of 1C18A via R117.
Resistor R125 provides a very small amount of
positive regenerative feedback to give clean square
wave switching. When the control voltage on pin
16B is O.OVDC, a square wave output swinging be¬
tween the negative rail and ground appears on the
output of IC18A. This voltage is divided down by
R131 and R132 to give a OVDC to -3.5VDC trian¬
gular wave output. When the voltage applied to the
control input on 16B is taken to -2.5VDC, a 15 per¬
cent duty cycle should appear on pin 1 of IC18A.
All of this paragraph is relevant to oscillators 1 and
2. However, oscillator 3 has a sawtooth inverter
circuit comprised of R123, R124, R128, R129,
R130, R133, R134 and Q20. This is a standard
common emitter transistor inverter with a biased
network on the output comprised of R133 and R134
to provide a reverse sawtooth signal which goes from
+1.75VDC to -1.75VDC (unloaded).
A minus 5 volt regulator comprised of the
voltage divider R9, RIO and emitter follower com¬
prised of IC2, Rll, R8 and Q1 provides a precision
-5 volt potential for the octave switching network
shown to the far left of the schematic. This circuit
has remote sense via pin 18A to eliminate effects
of edge connector voltage offsets.
Combined sawtooth and triangular waveforms
are achieved by the mix resistor shown to the right-
hand side of the schematic, off the printed circuit
board.
It should be noted that the voltage applied to
the base of exponentiating transistor Q17 is scaled
so that 21.0mV equals one octave. This means that
a 17.5 microvolt level on the base of Q17 is equiv¬
alent to 1 percent. If an accuracy of 1 percent is
required, then all noise contributions appearing at
the base of Q17 must be kept below 17.5 microvolts.
While this is not an unusually tight specification for
a 741 type operational amplifier, occasional problems
may be encountered. Popcorn noise or supply de¬
coupling in the integrated circuit or supply spiking
due to larger board trace resistances may lead to
occasional instruments which exhibit jitter.
2.4 MODULATION MIX AMPLIFIER
There are two modulation signals available in
the Minimoog; the output of Oscillator 3 and noise.
Oscillator 3 produces periodic modulation utilizing
triangle, sawtooth and pulse waveforms. Noise pro¬
duces random modulation utilizing noise voltages in
the pink and red spectrum. The Modulation Mix
amplifier selects either or both modulation signals,
sums them and routes them to the Modulation
Amount Control in the Left-hand controller. The
output of OSCILLATOR-3's WAVEFORM SELEC¬
TOR SWITCH, SW8, and the output of the NOISE
SELECTOR SWITCH. SW14, are fed thru R23 and
R24 respectively and to the MODULATION MIX
potentiometer R3. The wiper of R23 is connected to
ground and, therefore, when the MODULATION
MIX potentiometer is rotated, it pans between the
two modulation signals. The two ends of R3 feed
the input resistors of the Modulation Mix amplifier
composed of Q7, Q17 and Q18. The Modulation
Mix amplifier is an invertor with a gain of 2. The
output of the modulation mix amplifier is fed
through R57 to the AMOUNT of MODULATION
control in the Left-Hand Controller. Resistor R38
and capacitor C14 provide phase compensation to
the amplifier to suppress oscillation.
2.5 NOISE GENERATOR
The noise generator of the Minimoog uses a
small signal transistor operated in the avalanche
mode. The base-to-emitter junction is biased in
reverse breakdown which generates white noise in
the range of -60dB. This signal is amplified to pro¬
duce white, pink and red noise. The noise selector
switch selects white or pink noise for audio and pink
or red noise for modulation. Transistor Q15 is the
noise generator transistor which is selected, burned-
in and retested for uniform noise clear of pops and
2-5

clicks. Resistor R47 and capacitor C25 form a noise-
free bias supply for this transistor. The noise on Q15
is fed to common emitter amplifier Q12. Resistor
R26 adjusts the gain of this amplifier until the white
noise output is -5dB. Q4 forms an emitter follower
which provides a low impedance output drive for the
white noise circuit. The white noise output is filtered
by R16, C3, R8, C2 and R13 to provide pink noise.
Transistor Q3 amplifies the pink noise and provides
a low impedance output. The pink noise output is
then filtered by R12 and C7 and amplified by Q6 to
provide the red noise output.
2.6 HEADPHONE AMPLIFIER
The headphone amplifier in the Minimoog is a
push-pull current amplifier that provides sufficient
current to drive standard 8 ohm headphones. The
signal from the final VCA is fed to the headphone
volume control R21. This is AC coupled through
C19 through driver transistor Q16. Diodes, CR1 and
CR2 provide the 1.2 voltage drop required to bias
Q10 and Qll into class AB operation. CR1 and
CR2 are connected to Q10 and Q11 for temperature
compensation. Audio signals are AC coupled through
C21 into the headphone output jack.
2.7 POWER SUPPLY
The Minimoog employs a ± 10 volt dual regu¬
lated supply. The regulators are located on the left
hand portion of Board 3, the rectifier and filter
capacitors are located on Board 5, and the AC line
circuitry and low voltage transformer are mounted
on the chassis. The regulated supply can operate on
either 115 volt or 230 volt AC line voltage selected
by a rear panel line voltage switch. The supply has
both primary AC line fuses and secondary DC line
Raw AC from the line cord passes through fuse
F3 and to POWER switch SW20. VOLTAGE SELEC¬
TOR SW21 connects the primaries of T1 in series for
230 volt operation and in parallel for 115 volt opera¬
tion. The step-down secondary voltage from T1 is
applied to rectifier Board 5.
The low voltage AC is bridge rectified by diodes
CR1 through CR4 on Board 5. Capacitors Cl and
C2 filter the raw DC to provide unregulated +15
volts. This raw output is fed through secondary fuses
FI and F2 to Board 3. Raw +15 volts is also fed to
various boards in the system to provide additional
The power supply on the Minimoog employs
two series pass voltage regulators. The +10 volts is
regulated first and the -10 volts tracks the +10
voltage. The +10 voltage regulator is composed of
Ql, Q2, Q5, Q8, Q9 and referenced diode CR3.
Diode CR3 is the master voltage reference for the
-10 volt regulator and, therefore, the master refer¬
ence for the entire system. The negative voltage
regulator is composed of Q13, Q14, Q19 and Q20.
The +10 volt regulator operates in the following
manner: Resistor R44 feeds 7.5milliamperes through
zener diode CR3 giving a 6.2 voltage drop across
CR3. This 6.2 volts is virtually insensitive to both
changes in temperature and current, thereby provid¬
ing a very stable voltage reference. Transistors Q9
and Q8 compare the voltage drop across CR3 with a
voltage drop at the wiper of R21. To provide 6.2
volts at the wiper of R21, 10 volts must appear
across the total resistor string comprised of R39 and
R21 and R34. Therefore, the voltage across CR3 sets
the +10 volt power supply voltage. Transistors Q8,
Q9. Q5. and Ql form a negative feedback regulator
to maintain the voltage at exactly +10 volts. If the
voltage on the collector of Ql rises above 10 volts,
the voltage on the bases of Q8 ends up being lower
than the voltage on Q9. Transistor Q9 cuts off the
drive to Q5, which reduces the drive to Ql. This
brings the voltage on the collector of Ql back to +10
volts. Variable resistor R21 sets the output voltage to
precisely +10.000 volts. Transistor Q2 only operates
during the initial turn on of the instrument. When
the power is first applied, the collector of Ql has no
voltage on it, therefore the regulator will not func¬
tion. Transistor Q2 has no drive and is, therefore,
turned off allowing raw DC to be fed through R5
and R14 to the base of Q5, turning Q5 on. Transistor
Q5 supplies drive to Ql, turning Ql on causing the
voltage on the collector of Ql to rise towards +10
volts. As this voltage rises, Q2 is turned on, thereby
removing the raw DC feed to Q5 allowing the regula¬
tor to operate normally. Plus 10 volts is remote
sensed at the oscillator card to insure a stable oscil¬
lator voltage. Resistor R45 provides +10 volt sensing

in case the internal sense lines to the oscillator open.
Without R45, the regulated voltage would jump to
the raw supply voltage if the sense line opens. Resis¬
tor R45 protects the electronic circuitry from dam¬
age. The various capacitors throughout the system
suppress parasitic oscillations.
The -10 volt supply is referenced to the +10
voltage and the operation is identical to the +10 volt
supply. Resistors R65, R58 and R52 form a voltage
divider across the +10 and -10 volt supply rails.
Transistors Q13 and Q14 compare the voltage on the
wiper of R58 with ground potential. Transistors Q13
and Q14 adjust the drive to Q20 to maintain ground
potential on the wiper of R58 which provides -10
volts on the rail. Variable resistor R58 adjusts the
-10 voltage to precisely -10.000 volts. Again, remote
sensing is used for oscillator stability with sense
resistor R66 providing sense line protection and
various capacitors are placed throughout the regulator
to insure stable operation. Both the +10 volt and
-10 volt supply voltages are applied directly to the
oscillators and to Board 5 for distribution to the rest
of the system.
2.8 V OLTAGE CONTROLLED FILTER
Audio signals from the three VCO's, the noise
generator, and the external audio input are summed
and applied to the base of Q29. Transistors Q29 and
Q30 generate a differential signal current and apply
it to the lowpass filter ladder. The four-pole lowpass
filter ladder is a patented design consisting of the
base-emitter junction of transistors Q2, Q3, Q10,
Qll, Q19, Q20, Q23, Q24, and capacitors C16,
Cll, C7 and C3. Cutoff frequency of the filter is
controlled by the standing current through the lad¬
der. The higher the standing current, the higher the
cutoff frequency. The filtered signal is taken dif¬
ferentially across C3 and therefore the control
current is rejected. This differential signal is fed to
the gain recovery amplifier consisting of Q5, Q6, Q7,
and Q8 which converts the differential input to a
single ended output. This is then fed to Q16 in the
voltage control amplifier. The FILTER EMPHASIS
control, R14, is also connected to the gain recovery
amplifier to provide positive feedback at the cutoff
frequency. As the amount of emphasis is increased,
the Q increases until the filter breaks into a sine wave
oscillation at the cutoff frequency. The FILTER
EMPHASIS control is calibrated by resistor R73.
The cutoff frequency control current originates
in an linear voltage to exponential current convertor
consisting of a matched NPN-PNP pair Q26 and Q28.
Control voltages from the external input, the key¬
board switches, modulation, and the contour genera¬
tor are summed by their individual control resistors
and fed to the exponential convertor. Resistor R49
sets the keyboard scale at one volt per octave and
R39 calibrates the front panel filter CUTOFF FRE¬
QUENCY control.
2.9 VOLTAGE CONTROLLED AMPLIFIER
The input signal from the voltage-controlled
filter is AC coupled to the base of Q16. Transistors
Q16 and Q15 form a differential input transcon¬
ductance multiplier. The gain of the differential
amplifier is controlled by the current through Q18;
the greater the current, the higher the gain. Tran¬
sistor Q18 is connected to the loudness contour
generator which gives the signal from the VCF its
overall loudness contour. Resistor R14 adjusts the
common mode rejection of the first stage to mini¬
mize pops and clicks resulting from the control
voltage. The output of Q16 and Q15 is fed to another
transconductance multiplier consisting of Q14 and
Q13. The control voltage for this multiplier comes
from the EXTERNAL LOUDNESS control input
jack, J3. This shorting jack applies a positive bias
which keeps the VCA stage ON when external con¬
trol of loudness is not necessary. When an 1120 Foot
Pedal controller or another voltage controller is plug¬
ged into J3, this internal connection is broken and the
external applied control voltage controls loudness.
Resistor R12 balances this stage to minimize pops
and clicks. Transistors Q12 and Q17 convert the
differential signal to a signal-ended output voltage
which is routed to the main output and headphone
output VOLUME controls. The high level output
signal appears on J4 while a resistor divider consist¬
ing of R57 and R58 provide the low level output.
2.10 CONTOUR GENERATORS
The Minimoog has separate contour generators
for both filter and loudness contours. Each contour

generator has its own separate ATTACK, DECAY,
and SUSTAIN controls. The final decay rate is equal
to the initial decay rate and is selected by means of a
switch on the left hand controller. The contour
generators are triggered by an internal voltage trigger
from the keyboard and/or the rear panel S-trigger
jack. The filter contour generator consists of the
following parts:
• R-S Flip-Flop — Q1 and Q4
• Attack Time Electronic Switch — Q5
• Decay Time Electronic Switch — Q7
• Sustain Driver — Q8
• Voltage Follower — Q22 and Q21
• Trigger Driver — Q20
• Voltage Trigger Driver — Q12
• ATTACK TIME Control - R12
• DECAY TIME Control - R15
• SUSTAIN LEVEL Control - R18
• Amount of FILTER EMPHASIS
Control - R17
When there is no external S-trigger applied or
there are no keys depressed, the contour generator
is in the off mode. In this mode, transistor Q20 is
turned off and therefore, Q12 is turned on. Current
is fed through R37, R35, and CR1 to the base of Q4,
holding the flip-flop in the reset mode. Q12 shorts
CR2 and CR7 to ground eliminating the sustain
voltage and discharging C5 to ground. Therefore, the
output from pin 11 of the contour generator is
ground.
When any key on the keyboard is depressed, the
following events occur. The keyboard trigger on pin
20 of the contour board turns Q20 on. This removes
the reset voltage from the flip-flop and turns Q12
off removing the short on CR2 and CR7. The voltage
rise on the collector of Q12 is AC coupled to the
base of Q1 turning the flip-flop on. This turns Q5 on
putting +9.3 volts on the collector of Q5. Current is
then fed through the ATTACK TIME potentiometer
R12 to C5 resulting in an exponentially rising attack
voltage on C5. The resistance of R12 sets the RC
time constant and, therefore, the attack time. The
voltage on C5 is followed by Q22 and Q21 and
appears at one end of R33. When the voltage on R33
reaches approximately 5 volts, CR3 conducts, turn¬
ing on Q4 and resetting the flip-flop. This turns Q5
off and turns Q7 on. C5 is then discharged through
the decay control, R16, to the sustain voltage on the
emitter of Q8. The resistance of R16 sets this initial
decay time. The SUSTAIN LEVEL potentiometer,
R18, connected to pin 12, sets the voltage on Q8
which sets the sustain level. After the initial decay
time, C5 is maintained at the sustain level for as long
as a key is depressed.
When all keys are released, the contour genera¬
tor goes into the final decay mode. There are two
final decay modes available which are selected by the
final DECAY switch on the Left Hand Controller. In
the final DECAY ON position, CR7 is open circuited.
When all the keys are released, the emitter of Q8 is
shorted to ground and C5 is discharged through R15
resulting in a final decay constant rate to the initial
decay. In the final DECAY OFF position, CR7 is
coupled through a 1.5K resistor to the collector of
Q12. When all the keys are released, Q12 discharges
C5 abruptly resulting in no final decay. The output
from pin 11 is passed through the AMOUNT OF
CONTOUR control R17 and then to pin 16 on board
number 4. The AMOUNT OF CONTOUR control
adjusts the sweep of the voltage controlled low pass
filter.
The loudness contour generator is composed of
the following parts:
• Set Reset Flip-Flop — Q25 and Q15
• Attack Time Electronic Switch — Q16.R13
• ATTACK TLME Control - R13
• Decay Time Electronic Switch — Q13
• DECAY TIME Control - R16
2-8

• Voltage Follower — Q3 and Q2
• Sustain Driver — Q19
• SUSTAIN LEVEL Control - R19
The loudness contour section also utilizes Q20
and Q12, the trigger drivers and the final decay
switch. Operation of this circuit is identical to the
filter contour generator.
Both contour generators are driven from a de¬
coupled supply circuit which is derived from the +15
raw DC voltage. Since the timing capacitors can
generate 10 milliampere charging spikes, the de¬
coupling circuit keeps these spikes off the regulated
+ 10V. The base of Q26 is referenced to the +10
voltage so a +9.3 voltage follower appears at the
emitter. An external S-trigger is connected to the
base of Q12 through R42 for use with such acces¬
sories as the Moog SAMPLE and HOLD and RIB¬
BON CONTROLLER.
2.11 A-440 REFERENCE OSCILLATOR
The A-440 Reference Oscillator in the Minimoog
is based on a standard Wien Bridge oscillator design.
The Wien Bridge consists of the following parts;
R17, C13, R68, R71, C19, R55, and R50. The input
of the bridge is connected to the collector of Q9 and
the output of the bridge is connected to the base of
Q22, where it is amplified by Q22 and phase inverted
by Q9. At approximately 440Hz the Wien Bridge
has a phase shift of zero degrees, supplying positive
feedback, and the circuit will oscillate. The fre¬
quency of the oscillation is set by R68. Capacitors
CR3, CR4, C12, and resistor R27 limits the signal
amplitude at the collector of Q22 resulting in a clean
sine wave. Transistor Q4 buffers the output to reduce
circuit loading and provides a low impedance drive
to the VCA. The A-440 switch, SW18, activates the
A-440 reference oscillator by connecting the + rail
to +10 volts.
2.12 E XTERNAL PREAMPLIFIER AND
OVERLOAD LAMP DRIVER
The external preamplifier signal is fed into J6
to EXTERNAL INPUT VOLUME control R9. The
output of R9 is fed through R78 and C23 to the base
of Q27. Transistors Q27, Q32, and Q33 form a 200
gain amplifier. The output of Q33 is fed through
C20 to the EXTERNAL INPUT ON/OFF switch,
SW10. This switch allows an external source to be
summed with the other signal sources. The output of
Q33 is also fed to the overload lamp driver circuit.
Transistor Q25 detects the amplitude envelope of the
signal and feeds that to voltage divider R56 and R48.
Q33 follows this voltage and applies it to the base of
Q34. When the AC input voltage on Q25 reaches ap¬
proximately 1.2 volts, Q34 turns on lighting the
OVERLOAD lamp, Bl. R56 and R48 set the trigger
threshold and CR14 holds the OVERLOAD lamp on
for a suitable time period.
2.13 O LD MINIMOOG OSCILLATOR BOARD
(SERIAL NUMBERS BELOW 10175)
The Minimoog oscillator board contains three
independent voltage control oscillators which pro¬
duce sawtooth, triangle, and pulse wave forms. The
master TUNE control is provided to transpose the
entire instrument. Oscillators two and three have
individual tune controls for transposing individual
oscillators. Each oscillator also has its own independ¬
ent octave and waveform select switches. In addition,
oscillator three can also be used for modulation
purposes.
Z14 -5 VOLT REFERENCE SUPPLY
The -5 volt reference supply is derived from the
-10 volt rail and is used as a reference voltage on the
oscillator board. Resistors R173 and R177 divide the
-10 volt supply to -5 volts and apply that to the
buffer IC9 and Q38. The -5 volts appears at the
emitter of Q38 and is routed through the harness to
the octave switches and then back to the oscillator
board. The -5 volts is remote sensed at the octave
switches to compensate for connector resistance.
2.15 OCTAVE SWITCHES
Current from the -5 volt supply flows through
the octave resistors, R25 to R26, and then through
R169 and R168 to ground. Resistor R168 varies the
voltage drop across the octave resistors which adjust
the octave scaling. One voltage divider is shared by

all three octave switches. The output voltage from
each octave switch goes through its own separate
operational amplifier buffer to each one of the
oscillators.
2.16 OSCILLATOR ONE
Voltages from the pitch bend wheel, tune
potentiometer, keyboard, modulation wheel, exter¬
nal oscillator input, and the octave switch are
summed and inverted by IC1. This voltage is applied
to an exponential current convertor consisting of
transistors Qa, Qb, and Qc. The exponential con¬
vertor consists of two major parts; exponentiating
transistor, Qa, and a constant current reference
consisting of Qb, Qc, and IC3. As the input voltage
on IC1 increases, the output of 1C1 decreases, caus¬
ing the emitter voltage on Qb to decrease, in turn
causing the collector current of Qa to increase.
A one volt increase on the keyboard voltage input
causes a 20mV decrease at the output of IC1 which
generates a one octave increase in collector current
from Qa. This gives the oscillator the scale factor
of one volt per octave frequency change. The ex¬
ponential characteristic of Qa is not perfect since
there is a scale factor change with temperature.
TC resistor, R20, changes the gain of IC1 with
temperature to compensate for this effect. Also, be¬
cause of base resistance in the transistors, the expo¬
nential relationship tends to flatten at higher fre¬
quencies. Resistor R42 applies positive feedback
around IC1 to compensate for this effect. Resistor
Rll sets the frequency of the oscillator and R8 sets
the keyboard scale for one volt per octave. An RC
network across R78 phase compensates to maintain
AC stability.
The oscillator is a precision sawtooth relaxation
oscillator consisting of timing capacitor Cl, and 2 x
FET input amplifier consisting of Q7, Q8, Q9, Q4,
and Q3, a Schmitt trigger consisting of Q5 and Q6,
and discharge transistor Q10. The cycle starts with
Cl fully discharged and Q10 off. This leaves the
input of the FET amplifier at zero volts and the col¬
lector of Q3 at +4 volts. Transistor Qa pulls a con¬
stant current out of Cl causing a linearly decreasing
ramp voltage. When the voltage on Cl reaches -4
volts, the voltage on the collector of Q3 also reaches
-4 volts. This causes the Schmitt trigger, Q5 and
Q6 to fire, turning on transistor Q10 which resets Cl
to zero. Therefore, a +4 volt to -4 volt sawtooth
appears on the collector of Q3. The frequency of the
oscillator is controlled by the current coming from
Q3; the higher the cun-ent, the higher the frequency.
The amplitude of sawtooth generator is divided by
R33 and R34 and applied to the waveform switch.
The triangle wave is derived from the original
sawtooth by Q2. The +4 volts on the collector of Q3
causes the base-to-collector junction of Q2 to be for¬
ward biased resulting in +3.3 volts on the collector
of Q2. This voltage will follow sawtooth voltage
until the sawtooth passes zero volts. Then Q2
switches state causing the base-to-emitter junction to
forward bias forming an inverter. As the voltage on
the base of Q2 continues to drop, the voltage on the
collector of Q2 will rise, generating a triangle wave¬
form. Transistor Q1 buffers the output and sends
that to the waveform switch.
The variable duty-cycle pulse is also derived
from the sawtooth by Schmitt trigger transistors
Qll and Q12. The sawtooth is summed with a bias
voltage by R1 and R40. The bias voltage causes the
Schmitt trigger to fire at a specific point on the saw¬
tooth. Changing the bias voltage generates a variable
duty-cycle pulse. This pulse width is controlled by
the waveform switch and varies from a 50 percent to
a 15 percent duty cycle.
All waveforms are sent to the WAVEFORM
select switch, SW6, the LEVEL control R45, and
the OSCILLATOR ON OFF switch SW9. Resistors
R32, R33, and R34 supply the bias voltages for the
pulse width circuit. Resistors R36 and R37 on the
WAVEFORM select switch sum the sawtooth and
triangle to generate another waveform.
2.17 OSCILLATOR TWO
Oscillator two is identical to oscillator one with
one exception. Oscillator two has its own tune
potentiometer so that it can be varied in frequency
from oscillator one.
218 OSCILLATOR THREE
Oscillator three can be used as a tone oscil¬
lator or as a modulation oscillator. A switch, SW2,
2-10

interrupts the keyboard, modulation, external, and
pitchbend voltage on oscillator three and also in¬
creases the range of the oscillator three’s tune con¬
trol. This allows oscillator three to be used as a wide
range modulation oscillator. Transistor Q37 inverts
the sawtooth of oscillator three and applies that to
the WAVEFORM select switch so this waveform can
also be used for modulation.
2.19 KEYBOARD CIRCUIT
The keyboard circuit generates a low note
priority pitch voltage and a +10V (V-Trig) to ground
every time a key is depressed. The keyboard circuit
consists of three parts: a temperature compensated
current source, the actual keyboard, and the track
and hold amplifier.
Transistors Q9 and Qll form the temperature
compensated current source. Resistors R20 and R21
apply +5 volts to the base of Qll. Since Q9 and Qll
are complimentary matched NPN/PNP pairs, +5
volts appears at the emitter of Q9. This causes 8.5
milliamperes to flow through Rl, out the collector
of Q9, and through the 43 resistors in the keyboard
string to ground. This current flowing through the
10 ohm resistors in the keyboard string gives the
characteristic 1.0 volt per octave keyboard voltage.
Since the low end of the keyboard is grounded, the
keyboard voltage is the lowest note depressed giving
low note priority. The matched pair of transistors
result in temperature compensation and C3 stops
parasitic oscillations in the two transistors.
The keyboard voltage is routed to a track and
hold amplifier consisting of differential amplifier
Q23, Q24, and Q14, sampling switch Q13, storage
element C6, and output buffer Q10. With no keys
depressed, the circuit is in the “hold” mode. The
gate of Q13 is pulled down to -10 volts by CR5 and
R34, this turns Q13 off, C6 holds its “tracked”
voltage and Q10 follows this voltage. When any key
is depressed, two things happen: the pitch voltage
appears at the base of Q23 and +10 volts from the
trigger buss appears on the junction of CR5 and R34.
This reverse biases CR5, turning Q13 on, which
closes the amplifier feedback loop. Q24 then charges
or discharges C6 until the voltage on the source of
Q10 equals the input pitch voltage. Therefore, the
circuit has “tracked” the input voltage. Capacitor
C6’s charge time depends on the resistance of the
GLIDE potentiometer connected between pins 7
and 8 of the printed circuit board connector. With
no resistance (with the GLIDE switch off, for exam¬
ple) Q24 can charge C6 very rapidly resulting in an
instant pitch change. As the resistance is increased,
the charge time for C6 is increased resulting in the
familiar “Portamento” effect. When the key is
released, Q13 is turned off and the pitch voltage is
held. The keyboard is mechanically set up so that
the pitch voltage always occurs before the trigger and
that the trigger goes off before the pitch voltage is
released. This insures that the pitch voltage is con¬
stant when the amplifier goes from the “track to
hold” mode, and a reliable pitch voltage is held.
When no keys are depressed, R53 saturates the input
of the differential amplifier putting +10 volts on the
collector of Q14. This insures that Q13 will remain
pinched off, thereby holding the voltage on C6.
Capacitor C9 slows down the voltage on the input
of Q23 to minimize switch bounce noise.

PRINTED
CIRCUIT BOARD
SOCKET
KEYBOARD
(NOT VISIBLE)
FIGURE 2-3PRINTED CIRCUIT BOARD SOCKET LOCA TION