National Semiconductor LM380 User manual

TL/H/7380
LM380 Power Audio Amplifier AN-69
National Semiconductor
Application Note 69
December 1972
LM380 Power Audio
Amplifier
INTRODUCTION
The LM380 is a power audio amplifier intended for consum-
er applications. It features an internally fixed gain of 50
(34 dB) and an output which automatically centers itself at
one-half of the supply voltage. A unique input stage allows
inputs to be ground referenced or AC coupled as required.
The output stage of the LM380 is protected with both short
circuit current limiting and thermal shutdown circuitry. All of
these internally provided features result in a minimum exter-
nal parts count integrated circuit for audio applications.
This paper describes the circuit operation of the LM380, its
power handling capability, methods of volume and tone con-
trol, distortion, and various application circuits such as a
bridge amplifier, a power supply splitter, and a high input
impedance audio amplifier.
CIRCUIT DESCRIPTION
Figure 1 shows a simplified circuit schematic of the LM380.
The input stage is a PNP emitter-follower driving a PNP dif-
ferential pair with a slave current-source load. The PNP
input is chosen to reference the input to ground, thus en-
abling the input transducer to be directly coupled.
The output is biased to half the supply voltage by resistor
ratio R1/R2. Negative DC feedback, through resistor R2,
balances the differential stage with the output at half supply,
since R1e2R
2(
Figure 1 ).
The second stage is a common emitter voltage gain amplifi-
er with a current-source load. Internal compensation is pro-
vided by the pole-splitting capacitor CÊ. Pole-splitting com-
pensation is used to preserve wide power bandwidth
(100 kHz at 2W, 8X). The output is a quasi-complementary
pair emitter-follower.
The amplifier gain is internally fixed to 34 dB or 50. This is
accomplished by the internal feedback network R2–R3. The
gain is twice that of the ratio R2/R3due to the slave current-
source which provides the full differential gain of the input
stage.
TABLE I. Electrical Characteristics (Note 1)
Parameter Conditions Min Typ Max Units
Power Output (rms) 8Xloads, 3% T.H.D. (Notes 3,4) 2.5 Wrms
Gain 40 50 60 V/V
Output Voltage Swing 8Xload 14 Vp-p
Input Resistance 150k X
Total Harmonic Distortion Poe1W, (Notes4&5) 0.2 %
Power Supply Rejection Cbypass e5mF, ²e120 Hz 38 dB
(Note 2)
Supply Voltage Range 8 22 V
Bandwidth Poe2W, RLe8X100k Hz
Quiescent Output Voltage 8 9 10 V
Quiescent Supply Current 7 25 mA
Short Circuit Current 1.3 A
Note 1: VSe18V; TAe25§C unless otherwise specified.
Note 2: Rejection ratio referred to output.
Note 3: With device Pins 3, 4, 5, 10, 11, 12 soldered into a (/16×epoxy glass board with 2 ounce copper foil with a minimum surface of six square inches.
Note 4: If oscillation exists under some load conditions, add a 2.7Xresistor and 0.1 mF series network from Pin 8 to ground.
Note 5: Cbypass e0.47 mF on Pin 1.
Note 6: Pins 3, 4, 5, 10, 11, 12 at 50§C derates 25§C/W above 50§C case.
C1995 National Semiconductor Corporation RRD-B30M115/Printed in U. S. A.

TL/H/7380–1
FIGURE 1
GENERAL OPERATING CHARACTERISTICS
The output current of the LM380 is rated at 1.3A peak. The
14 pin dual-in-line package is rated at 35§C/W when sol-
dered into a printed circuit board with 6 square inches of 2
ounce copper foil (
Figure 2 ). Since the device junction tem-
perature is limited to 150§C via the thermal shutdown circuit-
ry, the package will support 3 watts dissipation at 50§C am-
bient or 3.7 watts at 25§C ambient.
Figure 2 shows the maximum package dissipation versus
ambient temperature for various amounts of heat sinking.
TL/H/7380–2
FIGURE 2. Device Dissipation vs Ambient Temperature
Figures 3a, b, and c show device dissipation versus output
power for various supply voltages and loads.
TL/H/7380–3
FIGURE 3a. Device Dissipation
vs Output Power Ð 4XLoad
TL/H/7380–4
FIGURE 3b. Device Dissipation
vs Output Power Ð 8XLoad
TL/H/7380–5
FIGURE 3c. Device Dissipation
vs Output Power Ð 16XLoad
2

The maximum device dissipation is obtained from Figure 2
for the heat sink and ambient temperature conditions under
which the device will be operating. With this maximum al-
lowed dissipation,
Figures 3a, b and c show the maximum
power supply allowed (to stay within dissipation limits) and
the output power delivered into 4, 8 or 16Xloads. The three
percent total-harmonic distortion line is approximately the
on-set of clipping.
TL/H/7380–6
FIGURE 4. Total Harmonic Distortion vs Frequency
Figure 4 shows total harmonic distortion versus frequency
for various output levels, while
Figure 5 shows the power
bandwidth of the LM380.
TL/H/7380–7
FIGURE 5. Output Voltage Gain vs Frequency
Power supply decoupling is achieved through the AC divider
formed by R1(Figure 1) and an external bypass capacitor.
Resistor R1is split into two 25 kXhalves providing a high
TL/H/7380–8
FIGURE 6. Supply Decoupling vs Frequency
source impedance for the integrator.
Figure 6 shows supply
decoupling versus frequency for various bypass capacitors.
BIASING
The simplified schematic of
Figure 1 shows that the LM380
is internally biased with the 150 kXresistance to ground.
This enables input transducers which are referenced to
ground to be direct-coupled to either the inverting or non-in-
verting inputs of the amplifier. The unused input may be
either: 1) left floating, 2) returned to ground through a resis-
tor or capacitor or 3) shorted to ground. In most applications
where the non-inverting input is used, the inverting input is
left floating. When the inverting input is used and the non-in-
verting input is left floating, the amplifier may be found to be
sensitive to board layout since stray coupling to the floating
input is positive feedback. This can be avoided by employ-
ing one of three alternatives: 1) AC grounding the unused
input with a small capacitor. This is preferred when using
high source impedance transducer. 2) Returning the unused
input to ground through a resistor. This is preferred when
using moderate to low DC source impedance transducers
and when output offset from half supply voltage is critical.
The resistor is made equal to the resistance of the input
transducer, thus maintaining balance in the input differential
amplifier and minimizing output offset. 3) Shorting the un-
used input to ground. This is used with low DC source im-
pedance transducers or when output offset voltage is non-
critical.
OSCILLATION
The normal power supply decoupling precautions should be
taken when installing the LM380. If VSis more than 2×to 3×
from the power supply filter capacitor it should be decou-
pled with a 0.1 mF disc ceramic capacitor at the VSterminal
of the IC.
The RCand CCshown as dotted line components on
Figure
7
and throughout this paper suppressesa5to10MHz
TL/H/7380–9
*For Stability With High Current Loads
FIGURE 7. Minimum Component Configuration
small amplitude oscillation which can occur during the nega-
tive swing into a load which draws high current. The oscilla-
tion is of course at too high of a frequency to pass through a
speaker, but it should be guarded against when operating in
an RF sensitive environment.
3

APPLICATIONS
With the internal biasing and compensation of the LM380,
the simplest and most basic circuit configuration requires
only an output coupling capacitor as seen in
Figure 7 .
An application of this basic configuration is the phonograph
amplifier where the addition of volume and tone controls is
required.
Figure 8 shows the LM380 with a voltage divider
volume control and high frequency roll-off tone control.
TL/H/7380–10
*For Stability with High Current Loads
FIGURE 8. Phono Amp
When maximum input impedance is required or the signal
attenuation of the voltage divider volume control is undesir-
able, a ‘‘common mode’’ volume control may be used as
seen in
Figure 9 .
TL/H/7380–11
*For Stability with High Current Loads
FIGURE 9. ‘‘Common Mode’’ Volume Control
With this volume control the source loading impedance is
only the input impedance of the amplifier when in the full-
volume position. This reduces to one-half the amplifier input
impedance at the zero volume position. Equation 1 de-
scribes the output voltage as a function of the potentiome-
ter setting.
VOUT e50 VIN #1b150 c103
k1RVa150 c103J0sk1s1(1)
TL/H/7380–12
*For Stability with High Current Loads
**Audio Tape Potentiometer (10% of RTat 50% Rotation)
FIGURE 10. ‘‘Common Mode’’ Volume and Tone Control
This ‘‘common mode’’ volume control can be combined with
a ‘‘common mode’’ tone control as seen in
Figure 10 .
This circuit has a distinct advantage over the circuit of
Fig-
ure 7
when transducers of high source impedance are used,
in that, the full input impedance of the amplifier is realized. It
also has an advantage with transducers of low source im-
pedance since the signal attenuation of the input voltage
divider is eliminated. The transfer function of the circuit of
Figure 10 is given by:
VOUT
VIN
e50 K1b150k
150ka
k1RTk2RVak2RV
j2qfc1
k1RTak2RVa1
j2qfc1L0sk1s1
0sk2s1
(2)
Figure 11 shows the response of the circuit of Figure 10 .
TL/H/7380–13
FIGURE 11. Tone Control Response
Most phonograph applications require frequency response
shaping to provide the RIAA equalization characteristic.
When recording, the low frequencies are attenuated to pre-
vent large undulations from destroying the record groove
walls. (Bass tones have higher energy content than high
frequency tones). Conversely, the high frequencies are em-
phasized to achieve greater signal-to-noise ratio. Therefore,
when played back the phono amplifier should have the in-
verse frequency response as shown in
Figure 12 .
TL/H/7380–14
FIGURE 12. RIAA Playback Equalization
This response is achieved with the circuit of Figure 13 .
The mid-band gain, between frequencies f2and f3, Figure
12
, is established by the ratio of R1to the input resistance
of the amplifier (150 kX).
4

Mid-band Gain eR1a150 kX
150 kX(3)
TL/H/7380–15
*For Stability with High Current Loads
FIGURE 13. RIAA Phono Amplifier
Capacitor C1sets the corner frequency f2where
R1eXC1.
C1e1
2qf2R1(4)
Capacitor C2establishes the corner frequency f3where XC2
equals the impedance of the inverting input. This is normally
150 kX. However, in the circuit of
Figure 13 negative feed-
back reduces the impedance at the inverting input as:
ZeZo
1aAob(5)
Where:
Zoeimpedance at node 6 without external feedback
(150 kX)
Aoegain without external feedback (50)
befeedback transfer function beAobA
AoA
Aeclosed loop gain with external feedback.
Therefore
C2e1
2qf3#Zo
1aAobJe1
2qf3#150k
1a50bJ(6)
BRIDGE AMPLIFIER
Where more power is desired than can be provided with one
amplifier, two amps may be used in the bridge configuration
shown in
Figure 14 .
TL/H/7380–16
*For Stability with High Current Loads
FIGURE 14. Bridge Configuration
This provides twice the voltage swing across the load for a
given supply, thereby, increasing the power capability by a
factor of four over the single amplifier. However, in most
cases the package dissipation will be the first parameter
limiting power delivered to the load. When this is the case,
the power capability of the bridge will be only twice that of
TL/H/7380–17
FIGURE 15A. 8XLoad
the single amplifier.
Figures 15A and B show output power
versus device package dissipation for both 8 and 16Xloads
in the bridge configuration. The 3% and 10% harmonic
TL/H/7380–18
FIGURE 15B. 16XLoad
distortion contours double back due to the thermal limiting
of the LM380. Different amounts of heat sinking will change
the point at which the distortion contours bend.
The quiescent output voltage of the LM380 is specified at 9
g1 volts with an 18 volt supply. Therefore, under the worst
case condition, it is possible to have two volts DC across
the load.
TL/H/7380–19
*For Stability with High Current Loads
FIGURE 16. Quiescent Balance Control
With an 8Xspeaker this 0.25A which may be excessive.
Three alternatives are available; 1) care can be taken to
match the quiescent voltages, 2) a non-polar capacitor may
be placed in series with the load, 3) the offset balance con-
trol of
Figure 16 may be used.
5

TL/H/7380–20
*For Stability with High Current Loads
FIGURE 17. Voltage Divider Input
TL/H/7380–21
*For Stability with High Current Loads
FIGURE 18. Intercom
The circuits of
Figures 14 and 16 employ the ‘‘common
mode’’ volume control as shown before. However, any of
the various input connection schemes discussed previously
may be used.
Figure 17 shows the bridge configuration with
the voltage divider input. As discussed in the ‘‘Biasing’’
section the undriven input may be AC or DC grounded. If VS
is an appreciable distance from the power supply (l3×) fil-
ter capacitor it should be decoupled with a 1 mF tantaulum
capacitor.
INTERCOM
The circuit of
Figure 18 provides a minimum component in-
tercom. With switch S1in the talk position, the speaker of
the master station acts as the microphone with the aid of
step-up transformer T1.
A turns ratio of 25 and a device gain of 50 allows a maxi-
mum loop gain of 1250. RVprovides a ‘‘common mode’’
volume control. Switching S1to the listen position reverses
the role of the master and remote speakers.
LOW COST DUAL SUPPLY
The circuit shown in
Figure 19 demonstrates a minimum
parts count method of symmetrically splitting a supply volt-
age. Unlike the normal R, C, and power zener diode tech-
TL/H/7380–22
FIGURE 19. Dual Supply
nique the LM380 circuit does not require a high standby
current and power dissipation to maintain regulation.
With a 20 volt input voltage (g10 volt output) the circuit
exhibits a change in output voltage of approximately 2% per
100 mA of unbalanced load change. Any balanced load
change will reflect only the regulation of the source voltage
VIN.
The theoretical plus and minus output tracking ability is
100% since the device will provide an output voltage at
one-half of the instantaneous supply voltage in the absence
of a capacitor on the bypass terminal. The actual error in
6

tracking will be directly proportional to the unbalance in the
quiescent output voltage. An optional potentiometer may be
placed at pin 1 as shown in
Figure 19 to null output offset.
The unbalanced current output for the circuit of
Figure 18 is
limited by the power dissipation of the package.
In the case of sustained unbalanced excess loads, the de-
vice will go into thermal limiting as the temperature sensing
circuit begins to function. For instantaneous high current
loads or short circuits the device limits the output current to
approximately 1.3 amperes until thermal shut-down takes
over or until the fault is removed.
HIGH INPUT IMPEDANCE CIRCUIT
The junction FET isolation circuit shown in
Figure 20 raises
the input impedance to 22 MXfor low frequency input sig-
nals. The gate to drain capacitance (2 pF maximum for the
KE4221 shown) of the FET limits the input impedance as
frequency increases.
TL/H/7380–23
FIGURE 20
At 20 kHz the reactance of this capacitor is approximately
bj4 MXgiving a net input impedance magnitude of 3.9 MX.
The values chosen for R1,R
2and C1provide an overall
circuit gain of at least 45 for the complete range of parame-
ters specified for the KE4221.
When using another FET device the relevant design equa-
tions are as follows:
AVe#R1
R1a1
gmJ(50) (7)
gmegm0 #1bVGS
VpJ(8)
VGS eIDSR1(9)
IDS eIDSS #1bVGS
VPJ2
(10)
The maximum value of R2is determined by the product of
the gate reverse leakage IGSS and R2. This voltage should
be 10 to 100 times smaller than VP. The output impedance
of the FET source follower is:
Roe1
gm(11)
so that the determining resistance for the interstage RC
time constant is the input resistance of the LM380.
BOOSTED GAIN USING POSITIVE FEEDBACK
For applications requiring gains higher than the internally
set gain of 50, it is possible to apply positive feedback
around the LM380 for closed loopgains of up to 300.
Figure
21
shows a practical example of an LM380 in a gain of 200
circuit.
TL/H/7380–24
FIGURE 21. Boosted Gain of 200
Using Positive Feedback
The equation describing the closed loop gain is:
AVCL ebAV(0)
1bAV(0)
1aR1
R2
(12)
where AV(0)is complex at high frequencies but is nominally
the 40 to 60 specified on the data sheet for the pass band
of the amplifier. If 1 aR1/R2approaches the value of
AV(0), the denominator of equation 12 approaches zero, the
closed loop gain increases toward infinity, and the circuit
oscillates. This is the reason for limiting the closed loop gain
values to 300 or less.
Figure 22 shows the loaded and un-
loaded bode plot for the circuit shown in
Figure 21 .
TL/H/7380–25
FIGURE 22. Boosted Gain Bode Plot
The 24 pF capacitor C2shown on
Figure 21 was added to
give an overdamped square wave response under full load
conditions. It causes a high frequency roll-off of:
f2e1
2qR2C2(13)
The circuit of
Figure 21 will have a very long (1000 sec) turn
on time if RLis not present, but only a 0.01 second turn on
time with an 8Xload.
7

AN-69 LM380 Power Audio Amplifier
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