Moog Micromoog 2090 Operating manual

TECHNICAL SERVICE MANUAL
for
MICROMOOG
and
MULTIMOOG
synthesizers
MICROMOOG
MODEL 2090
MULTIMOOG
MODEL 326A
James L. Scott • BSEE
(Author)
MOOG MUSIC INC.
2500 Walden Avenue
Buffalo. N*w York 14225
Copyright October. 1979

CONTENTS
SECTION
1
2
3
PAGE NO.
Title Page / Frontispiece .
Contents .
List of Illustrations .
List of Tables.
INTRODUCTION.
1.1 ORGANIZATION OF THE MANUAL
1.2 GENERAL SYSTEM DESCRlP“ION .
1.3 MODEL MODIFICATIONS.
CIRCUIT DESCRIPTION.
2.1 POWER SUPPLY.
2.2 KEYBOARD CIRCUIT.
2.2.1 DC Operation (Cortrol V/nlugo Generation)
22.2 AC Operation (Trigger Generation) .
2.3 CONTROL VOLTAGE SUMMEHS AM) DUAL
EXPONENTIAL CURRENT SOURCE . .
2.3.1 Oscillator Control Summing.
2.3.2 Filter Control Summing.
2.3.3 Dual Exponential Current Source .
2.4 OSCILLATOR. WAVESHAPER AND DOUBLER.
2 4.1 Oscillator.
2 4.2 Waveshaper.
2.4.3 Doubler.
2.5 VOLTAGE CONTROLLED IQWPASS FILTER .
2.6 VOLTAGE CONTROLLED AMPLIFIER.
2.7 DUAL CONTOUR GENEHATOP.
78 MODULATION.
2.9 NOISE SOURCE SAMPLE AND HOLD.
2.9.1 Noise Source.
2.9.2 Sample and Hold.
2.10 MISCELLANEOUSOFF BOARD COMPONENTS
Top Boord 2 .
Tandem Board 3 .
2-1
2-1
2 2
2-2
2-3
2-5
2-5
2-7
2-7
2-8
28
2-9
29
2-11
2-13
2-15
2 15
2-17
2-17
2-17
2-17
66
66
DISASSEMBLY PROCEDURES ...
3.1 OPENING THE SYNTHESIZER .. .
3 2 KEYBOARD REMOVAL.
3.3 MAIN PC BOARD REMOVAL ....
3.4 LEFT HAND CONTROLLER (LHC
3-1
3-1
35
35
3-10
TUNING AND ADJUSTMENT PROCEDURES.
4.1 EXTERNAL AND IN TfcHNAL ADJUSTMENTS.
4.2 EXTERNALLY ACCESSIBLE ADJUSTMENTS.
4.2.1 Oscillator Tuning.
4.2 1.1 Oscillator Range.
4.2 1.2 Oscillator Scale.
4.2.1.3 Oscillator High Frequency Compensation
4.2.1.4 OsciMotor Octavo Stop.
4-1
4-1
4-1
4 1
4 2
4-2
4 2
4 2
iv

CONTENTS (Continuod)
SECTION
4
5
4.3 FILTER TUMING.
4.3.1 Emphasis.
4.3.2 Filler Scalo.
4.3.3 Filter High Frequency Compensation.
4.3.4 Filter Octave Step.
4.4 OTHER ADJUSTMENTS..
4.4.1 VCA Balance.
4.4.2 Trill Trim.
4.4.3 Ribbon Centering.
4.5 INTERNALLY ACCESSIBLE ADJUSTMENTS.
4.6.1 - 16V ADJ.
4.5.2 Current Source Chip Temperature.
4 53 Filter Balance .
4 5 4 Filter Rang© ..
4.6 MULTIMOOG OSCILLATOR TUNING PROCEDURE .
4.6.1 Oscillator ' A" Temperature Adjustment . .
4.6.2 Oscillator "A" Tuning.
4.6.3 Interval Tuning Adjustment. .
TROUBLESHOOTING GUIDE.
5.1 SOUND CHAIN.
5.2 PITCH DRIFT AND TUNING.
b.2.1 Oscillator Drift.
6.2.2 O scillator and Filter.
5.3 KEYBOARD.
5.3.1 C leaning Key Contacts.
REPLACEMENT PARTS LIST.
MOTE: This lection also contains Multimoog circuit description and schematic diagrams
PAGE NO.
6 1
r

LIST OF ILLUSTRATIONS
FIGURE
1-1
1-2
1-3
1- 4
2 - 1
2-2
2-3
2-4
2-5
2-6
2-7
2-8
2- 9
2-10
3- 1
3-2
3 3
34
3-5
36
3-7
3-8
3 9
3-10
3-11
3-12
3-13
3 14
3- 15
4- 1
4-2
4-3
44
6 1
6-1
6-2
6 3
Rcor Panel, Early Version.
Rear Panel. Latest Version.
Control Panel, Early Version.
Control Panel, Latest Version.
Simplified Regulator Schematic . .. .
Keyboard Control Voltage Generation.
AC Equivalent of Keyboard String.
Equivalent AC Amp O/P Circuit.
Simplified Oscillator Circuit.
Waveshaper Output Shape Versus Waveshape Control Voltage.
VCF Frequency Response Curves tor Zero and Full Emphasis.
Equivalent Filter Ladder C rcuit.. • • • • ..
Envelope Shape (Either Section) at Timing Capacitor C705 or „706 .
Loudness Limiting.
Removing Case from Base.
Unplugging Audio Output Connector.
Unlocking Cable Clomp..
Base Assembly.
Installation of Grommet Strip.
Removing Ivor os.
Levelling Key.
Keyboard Mounting.
Servicing Switch Contacts.
Keyspring Replacement.
Knob Removal.
PC Board Removal.
PC Board In Service Position.
Left Hand Controller Removal.
Left Hand Controller Assembly, Part No. 93-603 .
Position for Making Externally Accessible Adjustments.
Oscillator Tuning Sound Chart.
Filter Tuning Sound Chart.
VCA Balance & Trill Trim Sound Chart.
Sound Chain Check .
Micromoog Block Diagram.
Micromoog Interconnect Harness Assembly.
Micromoog Schematic Diagram.
PAGE NO.
1-2
1-2
12
1-2
2-1
2-3
2 4
2-4
2-8
2-10
2-11
2-12
2-14
2- 15
3- 1
3-2
32
3-3
3-3
34
3-4
3 6
36
3-6
3-7
3-7
3-8
3-8
3- 9
4- 1
4- 2
4 3
4 4
5- 1
6- 3
6-5
6-7
LIST OF TABLES
TABLE
5-1 Sound Chain Troubleshooting Guide. .....
5 2 Oscillator and Filter Troubleshooting Guide
5-3 Keyboard Troubleshooting Guide.
5-4 Pots and Switches Troubleshooting Guide..
5-5 Control Circuits Troubleshoot ng Guide . ..
5-6 Power Supply Troubleshooting Guide ....
5-7 Noise, Etc. Troubleshooting Gu dc .
PAGE NO.
5-1
5-4
5-6
5-7
58
5-9
5-10
vi

SECTION 1
INTRODUCTION
This manual provides servicing and parts infor¬
mation for tho Micromoog/Multimoog Synthesizers.
11 ORGANIZATION OF THE MANUAL
The manual is divided into 6 sections. Section 1
provides general system description and model
modifications; Section 2 describes the circuit. Section
3 gives step-by-step procedures for disassembly of
the keyboard, main PC hoard and the left hand
controller. Section 4 describes the tuning and the
external and internal adjustment procedures. Section
5 contains the troubleshooting guide. To facilitate
ordering replacement parts, see .Section 6.
12 GENERAL SYSTEM DESCRIPTION
tSee Block Diagram — Figure 6-1)
The Synthesizers consist of several circuit blocks.
The POWER SUPPLY converts AC line voltage into
precisely regulated ±15 volts DC to operate all other
Micromoog circuits. A single VOLTAGE CON¬
TROLLED TONE OSCILLATOR (VCO) generates
the audio frequency waveforms that serve as the
source of pitched soundn. The wuveshaped output
of this oscillator passes through the VOLTAGE
CONTROLLED LOWPASS FILTER (VCF) which
strongly modifies the timbre ot the oscillator tone.
Tho NOISE SOURCE produces a pitchloa; hitting
sound that also may be fed into the filter. The outpu:
of the VCF pases through the VOLTAGE CON¬
TROLLED AMPLIFIER (VCA) to the Instrument
audio output. The* gain of the VCA is dynamically
shaped during a note by one section of the DUAL
CONTOUR GENERATOR. The VCA and contour
generator operating together constitute a loudness
ARTICULATOR The other section of the DUAL
CONTOUR GENERATOR dynamically shapes the
timbre of the VCF during a note. The contour
generator action is initiated by a trigger generated
within the KEYBOARD each time a key is depressed
The KEYBOARD also generates a control voltage
that increases as successively higher keys are depressed
This control voltage establishes the VCO pitch and
at the same time causes the VCF to track the VCO
to maintain constant timbre. The MODULATION
OSCILLATOR generates subaudio waveforms used
for control (not audio) purposes. This oscillator also
triggers the SAMPLE AND HOLD which samples the
noise circuit to yield control voltage that randomly
changes levels at a tempo sot by the RATE control.
The source of modulation is set by the SOURCE
switch and the circuit it will modulate is chosen by
the ROUTING switch. The MODULATION AMOUNT
wheel controls the level sent to the selected circuit
The Micromoog provides DC power to operate
auxiliary equipment. Through the rear panel the
Micromoog interfaces with external equipment such
as auxiliary controllers, other synthesizers, modular
equipment and other musical instruments.
1.3 MODEL MODIFICATIONS
This manual refers to the current production
model as of July 1976 - Serial Number 3300 and
above. Since the Micromoog was introduced to the
music world, several major and numerous minor
improvements have been incorporated. Some of
the modifications are discussed and illustrated in
tho following:
1) As of July 1976, the keyboard was modified.
It functions identically to and is interchangeable with
oirlior model::. Tho discroto rosUtorB and vortical PC
Board have been replaced by resistor packs and a
horizontal PC Board.
2) A direct audio output cable has been provided
to eliminate a slight, bleed thru. The cable is now
available as a field retrofit kit
3) A small rear panel PC Board and a control
pot (R261) has been added to provide variable
keyboard output control voltage. See Figures 1-1 and
1 - 2 .
4) As of October 1975, the nomenclature or the
main control panel. Serial Number 1500 and above,
was revised with no functional changes. See Figures
1-3 and 14.
1-1

FIGURE 1 1 REAR PANEL, EARLY VERSION
FIGURE 1-2 REAR PANEL, LATEST VERSION
FIGURE 13 CONTROL PANEL, EARLY VERSION
NEW
NOMENCLATURE
FIGURE I 4 CONTROL PANEL, t A TEST VERSION
1-2

SECTION 2
CIRCUIT DESCRIPTION
2.1 POWER SUPPLY 115 VAC. IC101 has been observed to survive the
1/4 second overvoltage that occurs at its inputs in
The supply druws 160 mA from a 115 VAC such ar. instance.
60 Hz input (or 75 mA from a 230 VAC 60 Hz input)
when the + 15V and - 15V outputs are 200 mA each.
Cl and C2 decouple RKI that may be riding in on the T1 steps the input AC down to 36 VAC cemer-
lino. SW1 connects the primaries of T1 in parallel for tapped The diode bridge CR101-CR104 rectifies the
115V operation and in series for 230V operation. FI AC to roughly ± 25 VDC. At a normal full load of
and 2 are in parallel for 115 VAC operation (250 mA ± 150 mA (Micromoog with 1125 Sample and Hold
blowout point) and in series for 230 VAC operation plugged into back), C101 and C102 provide sufficient
(125 mA blowout point). The fuses protect against filtering to keep the raw DC ripple around 2Vp-p. Dual
fire hazard should a short occur in the secondary of tracking regulator IC101 operates with external pass
Tl. They will not blow if the regulator outputs short transistors Q101 and Q102. These transistors drop the
out. (The legulator will current limit instead.) They ruw DC voltages to £ 15 DC. C103 and C104 stabilize
will blow i:' 230 VAC is applied and SW1 is set for the regulator against high frequency oscillation.
FIGURE 2-1 SIMPLIFIED REGULA TOR SCHEMA TIC
2-1

Since ihe base-emitter drop of external pass
transistor Q101 is about 0.6V, R101 will feed about
(0.6V/47 ohm - 12.5 mA) into IC101 pin 7, which
is more than sufficient to supply the 5 mA required
to operate internal chip circuitry. Qa passes the
balance of this 12.5 mA. Q101 forms n PNP-NPN
Darlington configuration with the internal pass
transistor Qa. Q101 passes mosl of the positive
current 1+ and therefore dissipates most of the heat
allowing IC101 to run cooler.
Pin 4 senses ♦ 15 V and causes the internal
circuits to drive Qa to maintain the output very
procisoly as the line and load vary. The sense input
provides another service; if 1+ reaches 250 mA, the
drop across R103 will rise to 0.7V and Ihe supply will
go into the current limit mode. If the output is
shorted to ground, Q101 will dissipate several times
normal power. However, Q101 will not burn out no
matter how long its output remains shorted due to
the attached heat sink. The negative regulator works
in exactly the same fashion as I he positive regulator.
Here are some important specs:
Input Voltage
Output Accuracy
Lcng Term Output Drift
Output Load Keg.
Output Lin* Reg
Output NoUe
Output Temp. Stab.
Output Current Limb Value
♦ 30V Max.
+ 10 mV (-1SV trimmed)
+ 300mV(+ 16 V ur.trimmedi
♦ 3 mV/1000 hours typical
30 mV Max. (0 to 230 mA)
20 mV Max. (SO VAC to
130 VAC 60Hz)
1.0 MV RMS Max. (100 Hz
to LOO kHz B.W.i
0.016%/°C Max.
250 mA
C105 and C104 are solid tantalum types to
effectively filter high frequency noise. CR105 and
CR106 protect the regulator against accidental con¬
nexion to it supply of an opposite polarity. RIOfi
sets the - 15V supply to - 15.00V. It must be se.
accurately since it is the reference for the Keyboard
String, Octave Buffer, Oscillator Range and Chip
Tomporaturo. The + 15V supply new! not he extremely
accurate so no trim is provided for it. It still must be
precise — i.e.: it must not drift or be noisy,especially
since the - 15V supply tracks the - 15V supply.
Cl09 and Cl 10 decouple the power supply lines on
tho main PC Board to forestall parasitic oscillations
The 1468 regulator is a ceramic encapsulated
device (non-hermetic glass frit type) rather than u
plastic device. The use of a plastic package would
result in a chip temperature of 125°C (not necessarily
worst case).This is dangerously close to the maximum
allowable chip temperature of 150°.High temperatures
cause shortened chip life, more rapid long-term drift,
and greater thermal (warm-up) drift. With ceramic
encapsulation the chip temperature is about 85°C -
a much more comfortable figure.
2.2 KEYBOARD CIRCUIT
Tho KEYBOARD CIRCUIT produces a control
voltage (to determine pitch) that increases linearly as
successively higher keys are depressed. The control
voltage increases from 0.00 volts at LO F (lowest, key)
at tho rate of one volt per octave (83.3 mV'semitone)
to a maximum of 2.58V at HI C (highest key 2-1/2
octaves above LO F). The keyboard has low-note
priority. If the performer holds down several keys
simultaneously, the lowest key determines the control
voltage.
The GLIDE control allows adjustment of the
control voltage transition rate from a 0.3 msec time
constant, (effectively instantaneous pitch change) to
a 1.0 second time constant, (effectively several seconds
before the pitch catches up to the controlling key and
stops changing). The Keyboard possesses a memory
that maintains tho control voltage at precisely the level
of the output at the instant the last key is released.
2.2.1 DC OPERATION (CONTROL VOLTAGE
GENERATION! (FIGURE 2-2)
IC202A applies constant current IK (set to
833 pA by R238) to the keyboard resistor string. The
1,0 F end of the string remains at ground potential
no matter how many or how few keys are closed.
Note that the I-R drop across the remainder of the
string below ihe lowest closed contact establishes
VOUT applied to the buss If more than one contact
closes, I K flows onto the bus at the highest closure
and returns to the string at the lowest closure. The
intervening resistors are simply shorted out.
2-2

FIGURE 22 KEYBOARD CONTROL VOLTAGE GENE RATION
When a key contact closes, holding capacitor
C212 charges (or discharges) at tho rate set by R251
to the level established on the buss. R250 sets a
minimum charging time constant of TCH (min) =0.22
pF x 1.5K “ 0.33 msec. R250 decouples the 25 kHz
pilot, oscillator signal (spp below) from the DC control
voltage on C212. R250 also filters key contact scrape
and bounce that occurs when a lower contact closes
while an upper contact is being held closed. When all
keys are released, no further charging of C2I2 is
possible and it holds the last instantaneous value of
control voltage. Dual J-FET Q201 in conjunction with
R252, R253 and IC204A comprise an extremely low
input bias current unity gain buffer. The buffer
provides substantial control voltage output drive
capability while draining C212 at less than a 200
pA rote.
MAX.MUN mSCHAROE ^ ^
NOTE
The output side of the buffer guards the
input side. This vastly reduces on-board
eakage since a potential equal to the value
being held surrounds tho entire holding
circuit.
C213 nnd C214 decouple RFI pickup by the
main PC Board. C214 also shunts capacitive audio
pickup by the keyboard control voltage output line.
CR203 and CR204 protect Q201 and 1C204A from
static discharge coming In via J3. R257 does not affect
the output li'vol normally (it. is inside the loop) but it
docs allow control voltage to be forced into J3 for
testing purposes, or for introducing external control
voltages to override the keyboard control.
22.2 AC OPERATION ITRIGGER GENERATION)
Since the keyboard uses a single-buss contact
scheme, nn AC pilot, signal (generated by IC201)
must he injected onto the buss so that detection of
contact opening and closing can occur. When closure
occurs, IC203 and IC202B amplify and detect the
signal and develop a clean DC trigger output.
Oscillator 1C201 generates the 25 kHz ± 5 kHz
triangular AC signal. 25 kHz lies above the audio
range which reduces the chance that unwanted signal
pickup by other Micromoog circuitry will be audible.
The 3080 transconduclance amplifier was chosen
as IC201 because it has very fas: slew rate, does not
delay coming out of input or output saturation and
does not draw current slugs from the supply lines
while slewing or while saturated. The output of a
3080 is a bilateral current source the magnitude of
which is given by:
lOUT _ 0-02 pA/ni» ( I npu t) . [ OCT (Mix.) - I BIAS
UA t bias)
R235 sets IFUAS (into pin 5) to 0.5 raa which
keeps the total package dissipation at 30 mW. well
below the absolute maximum of 125 rrW.lOUTwill
tend to alternate between + 0.5 mA and - 0.5 rnA,
2-3

FIGURE 2 3 AC EQUIVALENT OF KEYBOARD STRING
however the 50K impedance presented by K233/R234
will eauso tho iquarowav© output at pin 6 to saturate
at * 14 volts and - 14 volts instead. R234 and C201
integrate the squarewave into a triangular wave.
R232 and R233 set the triangular output swing to
± 1.25 volts. C204. R236 and R237 couple this signal
at a level of 15 mV peak (- 30 mV peak-to-peak)
to the non-inverting input of IC202A. C202 and C203
are the bypass capacitors (required for ever> 3080
amp) to keep IC201 from breaking into high
frequency oscillation.
Feedback via the keyboard resistor string force*
Vl to equal V 0. V2 supplied to t he buss is:
V2 - Vi (l "“ R ? - =>15 mV cV2 < 19mV
KX
This holds regardless of the number of keys
depressed or the setting of It Z.
It has been found that R239 does not completely
solvo tho marginal AC stability of IC202A. Tho
output of IC202A rings severely. Accordingly C215
has been added to bypass the equivalent R. L, C
impedance in the feedback loop presented by the
keyboard and its interconnecting cables.
Whenever one or more key contacts close, the
AC signal injected onto the buss couples via C206
and R240 to the input of IC203 The signal at the
input will exceed 6 mV peak even considering the
loading by R241, R242 and amplifier Z IN (worst
case - 10K). With IC203, another useful property of
the 3080 is exploited - the ability to provide very high
gain at ultrasonic frequencies.
I OUT (min. AC peak) - 6,7 x 6 mV - 40 uA
mv
R239 decouples IC202A from the keyboard Typically, 1 OUT will bo 100 pA peak to yield
buss capacitance when HI C closes to prevent the above equivalent output circuit (neglecting
IC202A from oscillating. CR206 for the moment).
FIGURE 2-4 EQUI VALENT AC AMP O/P CIHCUII
2-4

R244, R245 and R240 attenuate the peuk value
of V4 by a factor of 0.7, C210anri C211 attenuate if
another 0.7V, and the germanium rectifier CR201
drops it .300 mV. This yields a V5 of at least 700 mV
peak which easily overcomes the ' 200 mV threshold
of comparator IC202B set by R248 and R249.
Therefore, the output of IC202B snaps from - 14
volts (trigger off) to + 14 volts (trigger on). When the
trigger goej h:gh, R247 feeds 1 pA DC to R246
(whereas p-eviously it drained 1 pa). This current
dropped across R246 provides 200 mV of hysteresis
for IC202B.
When a key closes C211 charges quickly through
CR201(1 msec or less) but discharges slowly (10
msec) through R216. These lime constants in con¬
junction with the hysteresis given by positive feedback
resistor R247 provide excellent filtering of contact
bounce and scrape.
The circuit completely eliminates any bounce
with a period up to 10 msec. In addition, the 1 msec
charge up time rejects fast noise spikes. That is,
R245 assists in rejecting noise spikes whenever IC20.3
momentarily drives into saturation and sources 500 pA
(i.e., when Figure 2-4 equivalent circuit is no longer
valid).
R241 and R242 provide input bias current to
1020.3. At worst the DO input voltage due to VOs
and 1 OS will fall below 11 mV. R244 sets theoutput
DC level nominally to ground, however due to input
offsets the actual level may range from ' 8 VDC to
- 8 VDC. 0210 blocks this unwanted DC level.
C207 in conjunction with R244 rolls off :he output
of IC203 at 100 kHz to suppress oscillation. All
3080 amps operated in the linear modes must have a
similar capacitor.
When the performer releases all keys the AC
trigger signal cense* and the voltage nt pin 5 of IC
202B decays toward effectively + 100 mV (due to
hysteresis) until it crosses the + 200 mV threshold.
At that point, the output of IC202B switches back
to 14 volte and pin 5 proceeds buck to its rest vahio
of-100 mV.
As long as oscillator bleed through to the buss
stays below 1 mV peak, the AC output of IC203
will not be sufficient to misfire IC202B. At 20 kHz,
C212 has Xc = 1/(2 r fC) = 40 ohm. This attenuates
the trigger signal sneaking into the control voltage
output to at worst:
a 40 o a
19mVx IsoosT = 1/2mV
IC204 receives the external S-trigger input.
R235 end R256 set the input threshold to + 2 volts.
CR202 forms a logic OR function with CR207.
C213 and R254 debounce the external trigger. As
long as the external open-contact "on" time is less
than 15 msec, this circuit will bo offoctivo.
CR206 limits the input swing of IC202B to
+ 700 mV. if it were absent, whenever the output
of IC203 went high, C211 could charge to about
+ 6V. About. 5 time constants (50 msec) would be
required for the input of IC2023 :o decay to the
turn-off threshold after release of the external trigger.
The resultant 50 msec dead time between rotes is
unacceptable for keyboard performance A player
could play faster than the Micromoog could respond,
causing missed notes. CR20G cuts the discharge time
to roughly I or 2 time constants (10 to 20 msec) and
thereby prevents missed notes.
Newer models (after S/N 3300) have a small
auxiliary PC board at the rear above the power supply
board IC205 and the associated pot. R.2fil provide
an adjustable keyboard control voltage output of
from 0.90 to 1.10 volts per octave. Note that only
when the pot switch is in the “dick" (fully CCW)
position can the KRD CONTROI. OUTPUT jack
be used as an input. This arrangement allows a
“master" Micromoog to control a “slave" Micromoog.
The 11201 for the master can be used *s a scale tuner
for the slave if the slave R261 is clicked off.
2.3 CONTROL VOLTAGE SUMMERS AND DUAL
EXPONENTIAL CURRENT SOURCE
2.3.1 OSCILLATOR CONTROL SUMMING
The oscillator summer linearly adds the sevon
control voltages that influence oscillator frequency —
OCTAVE switch, FREQUENCY knob, FINE TUNE
knob, ribbon, modulation, EXTERNAL INPUT, and
keyboard. The weighted turn. Vo, feeds one input
2-5

of the dual exponential current source to produce
an oscillator control, Io, that doubles for each
increase of 20 mV applied to the base of Q301.
IO in turn sets the frequency ol the wide range
precision linear relaxation oscillator. Each doubling
of Io causes ;» one octave pitch rise The overall
effect yields a one octave pitch increase for each volt
increase of the- keyboard control voltage output.
IC301A and IC301B comprise an accurate
non inverting summing amp. the output of which rises
20 mV for each 10 uA fed into the summing node.
IC301A and IC301B operate at unity gain with
respect to the keyboard control voltage. R348 and
R349 attenuate the final output to a low eve!. This
minimizes the effects of RKI pickup and op-amp
noise and drift. R348 and R349 also prevont 1C30'B
from applying damaging voltages to IC304 during
instrument power-up.
Q307 and Q308 function as back-to-back Toner
diodes. Their leakage of .luA (max.) is far superior
to ’ real” zener diodes. They prevent IC301A from
saturating in response to extreme input drive into
the node b\ limiting the output summing of IC301A
to roughly ± 8V. If IC301A did saturate, the
oscillator control sum node (at pin 2) would deviate
from being u virtual ground. This would cause
oscillator control inputs Io couple via R330 and
R314 or via R372, R373 and R374 into the filler
control summing node (1C302A). The end result
would be a mysterious shift in filter frequency.
Q309 and Q310 similarly prevent the filter node
from influencing oscillator pitch.
It is possible for t.hc output of IC3GIB to rise
to - 8V (8 octaves above IO ■ 14 uA - See Dual
Exponential Current Source description that follows).
That is, 10 could rise to 3.2 m A. Thus current is apt to
overload current regulator 1C303A if Q304 and
Q305 are also conducting heavily. Accordingly R326
has been included in Ihe output circuit of IC301B
to limit the voltage applied to R348 to about ♦ 5V.
This limits Io to roughly 100 uA (40 kHz). CR303 is
included so as not :o limit the negative swing applied
to R348 and thereby limit the lowest oscillator
frequency obtainable.
OCTAVE switch SW301, divider chain R304,
R306, R307 and R309 and summing resistor R330
subtract 10 uA current increments from the oscillator
oonlrol node to shift the oscillator pitch clown in one
octave increments. R303, R305, R308 and R310
keep the divider chain outpul impedance constant
io prevent step inaccuracy due io loading. R329 trims
the step size. Since filter octave switching derives
from the same divider, the trim adjustments R329
and R311 will interact a little but not seriously.
FREQUENCY control R301 acts only when SW301
is set to “wide”. It sweeps the oscillator over a range
of * 4 octaves by applying current to the node
via R302.
When ROUTING switch SW802 rotates, the
input voltage offset (VlO) of IC301A will tend to
cause slight but annoying shifts in pitch. Resistors
R377 and R378 reduce this shift 90 7, by providing
a relatively low impedance path to ground for the
open contacts of SW802. Early models (S/N 1250
and lower, approximately) utilized an 1C selected for
low Vin for IC301A (VlO < 1/2 mV). Similarly
R378 and R379 prevent filter pitch shifting In earlier
models IC302A also was selected for 1/2 mV V IO.
SVV302 disconnects the keyboard output from
the oscillator in “drone” mode and it shorts out the
.mdio output of the waveshaper in “off” mode.
R335 and R336 set the oscillator froquoncy when all
inputs - 0 volts to 700 Hz. This occurs when
OCTAVE - 2' and keyboard LO F is depressed
(2* LO F) with all other inputs ofr. R334 and
R337 provide an adjustment range of ♦ 4 semitones
so “F” can be trimmed exactly to 698.5 Hz. Ribbon
R1003 with RI004 and R339 sweep the oscillator
over a range of + 8 semitones. R339 decouples the
oscillator summing node from th© cable harness.
CR1001 and CR1002 provide a deadband at the
center of the ribbon. R1001 and R1005 prevent the
ribbon from shorting + 15V to - 15V in the case both
ends of the ribbon are depressed simultaneously. 'Ilie
ribbon has no effect when not touched at all or when
touched exactly in the center. R1002 adjusts tile
electrical center to exactly coincide with the mechan¬
ical renter of the rihhon. R5 and R341 eive a rear
panel fine tune adjustment of + 2 semitones. 'Hie
response to an external control voltage input (J3)
is set to 13 semitones/volt by R343. R371 or R372
allows a nominal » 3 octaves frequency shift when
the modulation amount is turned up fully (triangle
modulation mode).
2-6

2.3.2 FILTER CONTROL SUMMING
IC302A and IC302B comprise a summer (just
like the oscillator summer) that adds the seven control
voltages that influonco filter cutoff frequency —
CUTOFF knob, OCTAVE switch, keyboard, external
input, modulation, filter contour and FILT. MOD.
BY OSC. switch. The filter control sum Vi- feeds the
other input of the current source to produce an
exponential output control current IF that sets the
cutoff frequency of the lovpass filter. A 10 uA
increase in the amount fed into the filter control
node causes VF to increase 20 mV and therefore
causes the cutoff to rise one octave (because IF
doubles).
Normally, the filter cutoff rises only 1/2 octave
for each volt rise in keyboard output. The filter
“half tracks” the oscillator to maintain constant
timbre up and down tho scale. However, when SW501
is set to “full” or “tone”, R324 parallels the normal
summing resistor R316 causing the filter to fully
track the oscillator. R321 adjusts the filter scale for
exact ono-to-one oscillator tracking.
Normally, OCTAVE switch SW301 moves the
filter cutoff in 1/2 octave steps. In the “norm." mode
a second section of SW501 shunts R312 to ground to
maintain a constant load on the octave divider chain.
In the “tone” or “full” mode, SW501 connects
R312 and R314 in parallel to yield full octave steps.
R311 adjusts the step size to exactly one octave.
CUTOFF control R320 tunes the filter over a + 5
octave range.
SW303 lets the tone oscillator “shake” the
filter (rapidly vary its cutoff frequency) to produce
useful timbral effects. R317 allows a “weak” shake
of a little over an octave. R322 allows a ‘ strong’'
shake of six octaves. The response to external
control voltage input (J4) is 13 semitones/volt asset
by R31H. R38U and R381 provide a i 1 octave range
adjustment so that when ill control inputs equal zero
volts, the filter cutoff is 700 Hz a 2' LO F.
When modulation switch SW802 is set to “osc
and filt”, the filter normally half-tracks the oscillator
However, if the performer sets SW501 to “lone” or
“full”, R373 parallels R374 causing the filter to track
the oscillator exactly. The “filt'’ position of SW802
allows a filter deviation oft 6 octaves with the modu¬
lation wheel turned up fully (triangular modulation).
2.3.3 DUAL EXPONENTIAL CURRENT SOURCE
Q301, 302, 303, 304 and 305 are contained
on a single 1C chip mounted in a standard 14 lead
plastic DIP. As Io, II' and IF'vary, IC303A draws
in current through CR301 to maintain the collector
of Q302 exactly steady at Vk d + 750 mV (+ 250 mV
depending on IC303A input bias current). This
st&bilizos the reference current IR which flows through
Q302 to 14 pA. Oscillator current source Q301
balances against reference transistor Q302 to produce
a highly precise current given by:
IO = IR e< Vo;VT >
. A kT
where: VT - -
q
and: AVT/ AT - +0.33%/ °C
k - Boltzmann’* constant
T ■ absolute Kelvin temperature
q - charge of one electron
R364 and R365 establish Vft at + 750 mV. The
primary purpose of this is to keep the input bias
current of IC303A from dragging the collector of
Q302 below ground potential, thereby spoiling its
operation. CR301 prevents positive excursions of tho
output of IC303A from damaging IC304. R366 and
C307 stabilize IC303A against oscillation by reducing
its AC gain by 6(JdB above 1.5 kHz. C303 and C3U4
squelch parasitic oscillation and RFI pick jp.
At the design chip temperature of 55° C. Vo
must change exactly 19.55 mV to cause IO to double
(cr halve). This value of AVo increases 0.0607 mV/°C.
In order to maintain the scale factor AVo precisely
constant, T (chip absolute temperature) must be
carefully regulated. Now since a precision reference
current IR passes through Q302 VF precisely
measures temperature. AVE/AT is about -1.9
mV/°C. Accordingly, we make Q302 serve two pur¬
poses: that of thermometer as well as reference device.
R358, R359 and R360 establish a temperature
“set” potential Vs of approximately - 0.55 volt. As
the temperature of the chip decreases, the base*
2-7

emitter potential nf Q302 increases. This results in
VE becoiring more negative. If VE drops below Vs,
the output of IC303B goes negative. This output is
applied via emitter follower Q306 to the bottom
of R328. The potential at the top of R.128 is fixed
at + 1 volt by the emitter follower consisting of
Q303 and bias resistors R367 and R368. Thus, the
potential applied to the bottom of R328 sets the
current that passes through the chip heater. Q303.
Since the C-E potential is always about 14 volts,
Q303 develops a maximum of 14 volts x 28 m.A =
400 mW. As soon as tho chip temporaturo rises
enough so that Vl - Vs, the output of IC303B goes
more positive to reduce the current drawn through
Q303 and thereby maintain constant chip temperature.
Bias resistors R367 and R368 prevent, the emitter
of Q303 from going below the substrate potential
(pin 13 is connected to the substrate). Also this biasing
keeps the C-F. potential below thp absolute chip
maximum of 15V. A potential of + IV here also
allows complete shut off of the heater thanks to the
junction drops of Q308 and CR302. Q306 sinks more
current than a 1458 op amp alone could provide.
Also, use of Q306 avoids self healing of IC303B
(and IC303A. since it is on the same chip). Such
heating would upset VOS of both of these devices
and contribute to IR and T errors. CR302 and R362
prevent breakdown ol the base of Q306 should the
output of IC303B rise above ♦ 5 volts.
The chip-to-ambient thermal resistance 4>JA
of the current source was measured as 140°C/watt
(RCA plastic CA3046 in DIP socket). This implies
thiil the maximum temporaturo rise achievable is
about 55°C (140°C/watt x .4 watt) and therefore
that the Micromoog will operate in ambients down
to 0°C. In a 25°C ambient Q303 will dissipate 140 mW
which Implies that Its current is 10 mA. Under normal ' '
conditions tho chip temperature reaches 55°C in
about 5 minutes. After this time a few more minutes
are required for thermal gradients to settle out in the
body and leads of the CA304G. Consequently, a total
of about 10 minutes is required before the 10 and IF
scale factors to become rock-solid.
Filter current source transistor Q305also balances
against reference transistor 0302. Filter control
current IF is produced inexactly the same fashion as
IO. IF’ will be explained in Section 5.
2.4 OSCILLATOR. WAVESHAPER AND
DOUBLER
2.4.1 OSCILLATOR
10 steadily discharges C401 to produce a linearly
declining ramp. The FET input amplifier consisting
of Q402, R406, R407 and IC401B provides extremely ^
low input current unity gain buffering of the capacitor
ramp voltage. When the ramp reaches zero volts, the
trigger circuit. (R408. R409. R410. R411. R412.
R413.C403, C404, C405, C406, C407, IC402, CR401)
generates a 6 psec positive pulse that turns on switch
Q401 momentarily. Q401 shorts out C401 causing the
sawtooth wave to reset up to the + 10V reference
(produced by R403, R404, R405, C402, C410, and
IC401A). C401 is a polystyrene capacitor for good
AC characteristics, good stability, and low oakago.
f
2-8

R408 and R409 attenuate the sawtooth applied
to IC402 so as not to exceed its differential input
limit. R412 feeds a high value of amplifier bias
current (1mA ) to IC402 to maximize its speed. It
would seem that this would pul IC402 dissipation
at 100 mW (dangerously close to the package limit
of 125 mW). However, since the average output
current is nearly zero, the dissipation is exactly
halved to 50 mW.
The network consisting of C405, R410 and
R411 feeds hack a positive pulse (1.5 psec time
constant) lo the plus input of IC402.This precludes
latchup of the reset loop. C406 stabilizes IC402 to
prevent high frequency oscillation. In addition it limits
the output risetime to 1 psec This risetime in
conjunction with the 1.5 psec time constant yield a
pulse height a: pin 3 of roughly 2.5 volts instead of
the expected 5 volts. Thus, the pulse slays within
the differential input limit of IC402.
C407 speeds up Q401 turn-on. C402 and C410
steady the * 10 volt reference. Pull up resistor R413
allows the gate of Q401 to float to the source
potential whenCR401 us back-biased (3080pulse high).
High frcquoncy compensation resistors R401
and R402 correct for the dead time of the integrating
capacitor. This time equals the width of the reset
pulse (4 to 8 psec). The theoretical value for R40i
plus R102 = reset time + integrating capacitor =
(6 x 10-6) 4 (IQ-®) - 6K. The amount of
compensation :n cycles per second is directly propor¬
tional to lo (and oscillator frequency).The compen¬
sation counteract* “high end droop” It lowers the
voltage reached by the positive peak of the sawtooth
wave by an amount equa to lo x (R401 x R402).
The integration lime of the declining ramp is reduced,
thereby boosting the frequency.
2.42 WAVESHAPER
This voltage controlled circuit shapes the wave¬
form continuously from sawtooth through square
and on to narrow pulse. As R414 sweeps from CCW
to CW the output of IC403A (Vw) declines from
0V to - 10V. At 40% electrical rotation, the sawtooth
ware has been totally truncated. At 100% rotation,
the output reduces to a very narrow pulse (- 10 usee).
R419 and C409 act to prevent the output from
“narrowing to nothing” by putting a spike on the
positive swing of the sawtooth wave fed to Q404.
R416 allows a modulation swoop roughly oquivalont to
100% pot rotation (triangular modulation waveform).
R421, R422, K423 and Q403 constitute the
sawtooth truncator. When Vw roaches - 4 volts,
Q403 and R421 gobble up all the current that R422
can supply, leaving no signal for summing resistor
R423 Lo feed lo signal adder IC403B.
The rectangular wave duty cycle depends on the
zero crossing times of the level-shifted sawtooth seen
at the base of Q404. When Vw --5V,the duty cycle
= 50%. etr. R437 offsets the level shift slightly so that
when Vw - 0, there will be no pulse output, and the
sawtooth waveform will be pure.
R424, R425, R426. Q404, Q405 and CR402
comprise a bi-stable zero crossing detector (Schmitt
trigger circuit). K424 and CK402 set the common
emitter voltage ut - 0.7V regardless of the state of
Q404 and Q405. This establishes the threshold point
of the Schmitt at approximately ground (+ 50 mV).
Circuit regeneration produces a dean snappy output
and the input hysteresis is about 10 mV. Summing
resistor R429 feeds the adder IC403B. The AC
level of either the pure sawtooth wave or the pure
rectangular wave is 2 volts p-p at the output (pin 7).
2.4.3 DOUBLEK
The doubler produces exactly 50% duty cycle
square waves one octave and two octaves below the
primary oscillator pitch. Q406 inverts and level shifts
lh« oscillator reset pulse. The rising edge of this
inverted pulse clocks the input of IC404 Section A
(connected as a toggle fiip-flop). The output alter¬
nately transists from + ■ 15V to 0V and then from OV
to + 15V on the primary sawtooth wave reset pulse.
F/F A feeds an identical stage, F/F B. F/F B
transitions occur on the positive excursions of F/F A.
R431, R432, and R436 attenuate and level shift the
octave 1 signal. R433, R434 and R438 do exactly
the same to the octave 2 signal.
The octave signals feed the ends of R435. The
waveshaped primary signal feeds the center tap. With
2-9

the control knob centered only the pare primary
signal feeds cut the wiper. As the control rotates to
either hide of center, the primary signal and cither
octave I or octave 2 is mixed with the primary in
such a fashion that the summed output level stays
approximately constant. At the extremes of rotation,
the output consists of about 60% octuvo and -10%
primurv (peak-to-peak voltage basis).
One might reasonably ask why R436 and R438
are required lo level shift the octave signals down so
that their I>0 level in roughly ground. Aftor all. C506
AC couples the mixed signal into the VCF so the DC l
shift seems to be unnecessary. However, if R436 and
K438 are eliminated, and if one then routes the
DOUBLING knob rapidly back and forth, the DC
level at the wiper of R435 will suffer large shifts.
IC403B O/P WAVEFORM
ro- 10.0 V
COMMENTS
PURE SAWTOOTH
0 % RECT. DUTY CYCLE
-3 VDC OFFSET
RECTANGULAR WAVE APPEARS
ONSET OF
SAWTOOTH
TRUNCATION
SAWTOOTH DISAPPEARS
RFCT DUTY CYCLE ~40%
50% RECT. DUTY CYCLE
- 1 VDC OFFSET
HECT. DUTY CYCLE ^90%
{NARROW PULSF)
FIGURE 2 6 WA VESHAPER OUTPUT SHAPE VERSUS WA VESHAPE CONTROL VOL TAGE
210

C506 will couple the transients tr> the VCF. momen¬
tarily causing the filter ladder to go out of balance
and to produce a harsh “gritch" sound. Resistors
R436 one R438 alleviate the problem. Also, C506
Is chosen so that the coupling time constant into
the VCF is about 25 msec (6 Hz) to further preclude
difficulties.
2.5 VOLTAGE CONTROLLED LOWPASS FILTER
The cutoff frequency of the so-called VCF is
actually current controlled by IF. The VCF attenuates
high frequency components of the signal while
allowing low frequency components to pass through.
An emphasis circuit can be brought into play (R525)
to peek up the response at the cutoff point. When
this is done, the pa»>band level will be uniformly
depressed. In the ‘‘tone” mode, the emphasis is
increased to the point that the filler breaks into
oscillation and acts as another tone source accurately
playable from 40 Hz to 4 kHz. (See Figure 2-7)
The filter is a refined version of the hUindard
Moog ladder circuit. R502. R503 and R511 attenuate
the oscillator output to a low level signal. This AC
signal (nominal sawtooth level - 40 mV p-p) feeds to
differential current driver pair Q501 and Q502 at the
bottom of the ladder. Q501 and Q502 n turn drive
the four filter sections of the ladder (stacked one atop
the other). R005 through R510 bias each pair of
transistors in the ladder 1,5Vabove t he preceding stage.
C514 and C506 block input DC. C507 bypasses
AC signal around R510. C508 and C505 suppress
RFI pickup. R501 adds the external audio signal to
the VCF input. 100 mV RMS drives the Micro to full
output (+12dBm>. R504 adds pink noise to the VCF
input. R513 limils IF to about 1.2 mA (limits fc to
60 kHz) so as not to overload Q305 It also protects
the ladder from accidental burnout. Note that when
SW302 turns the tone oscillator off, it does so by
killing the audio feed to the VCF while leaving the
oscillator running. This scheme retains capability
for filter modulation by the tone oscillator (via
C412) when the OSCILLATOR is -off". However,
note that when both SW302 and SW303 aro off,
the oscillator is killed by shorting pin 3 of IC402
to ground. This feature (instituted in newer models
S/N 3300 and above) prevents the VCO from
bleeding into the VCF when the VCF is acting as a
pure sinewave tone source. The bleed would cause
generation of impure YCFsinewave tones at high pitch.
FIGURE 2 7 VCF FREQUENC Y RESPONSE CUR VES FOR ZERO AND FUL L EMPHASIS
2-11

Q503, Q504 and C501 constitute the first rung
of the ladder. The three Identical sections above it
constitute the other three rungs. Q501 and Q502 split
the control current IF approximately evenly .One-half
IF flows down the left side through Q5O3,Q505, Q507
and Q509 while the other half flows down the right
side through Q504, Q506, Q508 and QBlO.Now the
incremental base-emitter junction resistance of a
ladder transhtor is given by:
RE = where IC « 1/21F
<g> 1C - 1 mA, HE - 25 il
Since the bases of Q503 and Q504 are connected
in series. C501 sees an effective resistance given by:
RP - 2 RE -
100 ft
mA
100K
u'A~
The cutoff frequency for the stage is then:
2 ir RF C
IP -
50 Hz
/uA
At fc the attenuation is 3 dB and the phase shift
is 45° for a single rung. Since the collectors of Q503
and Q504 feed the next rung, there is no interaction
between these stages or any other stages. We have in
effect the following AC circuit. (See Figure 2-8).
This is a four section turnable RC circuit with a
cutoff slope of 24dB/octave, 12dB attenuation at fc
and 180° phase shift at fc.
Note that Q501 and Q502 is not really the
idealized linear driver as shown in Figure 2-8. It is
actually slightly exponential. However, when tne
filter is “wide open** (fc - 60kHz), the audio
frequency input signal is relatively undistorted at
the output since Q509 and Q510 present a logarithmic
load characteristic. This compensation causes the
differential output (seen across the emitters of QB06
and Q510) to have the same level as and to have the
same waveshape as the input applied to Q501. Q501.
Q502, Q509 and Q510 must be matched and must
track well. They are, therefore, contained on a
single 1C transistor array so the ladder remains
properly balanced. The parameters o: the other
transistors in the ladder are not critical (and these
components are not matched).
The input level was chosen rather high at 10 mV
(p-p) for good signal to noise (S/N) ratio. The high
level drive results in appreciable harmonic distortion,
but this is harmless (if not beneficial). There is, of
course, no intermodulation distortion to worry about
in a one oscillator Instrument. The level could even be
higher if we didn’t have to worry about excessive
overdrive masking the effects of emphasis.
Q514 and IC501 comprise a differential input,
single sided output, gain recovery amplifier. Dual
4-FFT Q514 acts as a dual source follower to keep
1C501 from drawing input bias current directly from
the top of the filter ladder. If permitted, this drain
would throw rhe ladder severely out cf tune at low
frequencies. R517 sets IC501 amplifier bias current
(1 ABC) into pin 5 at 500 uA, about the highest level
practical. A lower value of (I ABC (would dogonorace
the S/N ratio of the gain recovery amp and also
impair its frequency response at the high end of the
audio spectrum. R518 and R519 set the output DC
level of IC501 to + CV and R517, R518 and R519 w>t
FIGURE 2 8 EQUIVALENT Fit TER LADDER CIRCUIT

its AC gain to -» 40dB. A Darlington follower (Q512,
Q513. and R521) buffers the output to provide low
impedance drive to the emphasis circuit. R527 pro¬
tects the buffer transistors from accidental burnout
should the emitter of Q5I3 be shorted to ground.
The emphasis network feeds hack to ladder driver
Q502. In the passband, the feedback subtracts from
the input signal (180° phase shift) causing the overall
level to drop. At the cutoff frequency, however, the
180° phase reversal contributed by the ladder results
in a net shift of 0 ° (positive feedback) and a regen¬
erative peak appears in the frequency response curve -
i.e., "emphasis".
Now here is what we want. In the “norm" and
‘‘full” modes, the filter MUST NOT quite oscillate at
any fc with the emphasis knob (R525) fully up. This
will be true if the loop gain remains slightly less than
unity.
In the "normal" and "full" modes of SW510,
R522 and R521, attenuate the feedback 26dB. Now
with R525 - 0. we adjust R526 for a little more loss
(about 2dB) and the overall loop gain nears unity.
♦ 40dB (amplifier) - 12 dB (chain) - 26 dB (emphasis
feed) - 2 dB (trim) - 0 dB. Actually R526 is set so
the loop gain - - 0.5dB. This yields a strongly
resonant peak with Q of-10 max (threshold of
oscillation).
New here is the other constraint: in the “tone”
mode, the filter MUST oscillate (act as a sir.ewave
tone source) from 20 Hz to 10 kHz. In the “tone”
mode, R523 instead of R522 feeds the emphasis
network which boosts the loop gain to + 2.5 dB.
Secure and stable oscillation results. In the “tone”
mode SW501 shorts R525 so that oscillation will
occur no matter where the EMPHASIS knob is set.
During oscillation the ladder self limits at about 40 mV
p-p resulting in the same output level one would
obtain from the VCO. The output is a quite pure
sinewave that is very accurate when played from
the keyboard (over the full range of the synthesizer).
If the VCO audio feed to the filter remains on while
*ho filter produces its own tone, we obtain a very
nice effect. When the filter frequency is near the tone
oscillator frequency (or a harmonic of the oscillator
frequency),ihe filter will lock onto the tone oscillator.
Nominally fc should exactly double each time
VF (hasp of Q305) rises 19 55 mV However, at
higher frequencies fc does not rise quite high enough
(“high and droop") causing the pitch of the oscillating
filter to go flat in the X and 2' ranges. It is found that
to compensate for the error we need to boost the
pitch by an amount; proportional to fc.
To compensate properly, IC304 derives positive
feedback from otherwise spare transistor Q304. Q304
parallels Q305 so that IF' - IF. At 1 kHz, R369
develops about 4 mV which feeds at a gain of
roughly 2 through IC302B. After sttcnuution by
R354 and R355, this signal amounts to about
.2mV (1/100 of an octave or 20 cents). R357 trims
this corrective feedback. 1(376 limits IF'so as not to
overload Q304 or 1C303A. R376 also provents
danger of latch up of IC302B due to excessive
positive feedback.
2.6 VOLTAGE CONTROLLED AMPLIFIER
Here the versatile 3380 is found in another
application — that of a variable gain amplifier. One
section of the Dual Contour Generator produces an
envelope buffered by IC602A. As the voltage envelope
swings to its maximum value of + 9V, R608
establishes a current proportional to the envelope
voltage with a peak value of 500 uA which flows
through lewd shifter Q601. This bias current passes
into IC601 pin 5 to control its gain. With no bias
curront flowing, IC601 shuts off completely. Other¬
wise, the signal gain is proportional to the bias current.
The signal output current of IC601 passes to the
AC current-to-voltage converter comprised of R609
and IC602B. IC602B acts as the output buffer and
drives VOLUME pot R612. Protective devices R611,
CR601 and CR602 prevent external static discharges
or chassis potential differences from damagingIC602B.
C605 keeps DC out of the output and C604
suppresses oscillation. RG13 supplies a DC output
ground reference for IC601. R614 provides sensing
so that fault, isolation can be accomplished at the
output of 1C601. When "On”, SW601 keeps the VC A
fully on regardless of the envelope voltage. R607
limits the drive to IC601 to 500 pA (0601 will
saturate) during bypass or when the articulator
contour generator is in the sustain mode.
2-13