Linear Technology LTC1872 User manual

LTC1872
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For more information www.linear.com/LTC1872
Typical applicaTion
FeaTures DescripTion
Constant Frequency
Current Mode Step-Up
DC/DC Controller in SOT-23
The LT C
®
1872 is a constant frequency current mode step-
up DC/DC controller providing excellent AC and DC load
and line regulation.The device incorporates an accurate
undervoltagelockoutfeaturethatshutsdown the LTC1872
when the input voltage falls below 2.0V.
The LTC1872 boasts a ±2.5% output voltage accuracy
and consumes only 270µA of quiescent current.For ap-
plications where efficiency is a prime consideration,the
LTC1872 is configured for Burst Mode operation,which
enhances efficiency at low output current.
In shutdown,the device draws a mere 8µA. The high
550kHz constant operating frequency allows the use of a
small external inductor.
The LTC1872 is available in a small footprint 6-lead
SOT-23.
applicaTions
n High Efficiency: Over 90%
n High Output Currents Easily Achieved
n Wide VIN Range: 2.5V to 9.8V
n VOUT Limited Only by External Components
n Constant Frequency 550kHz Operation
n Burst Mode™ Operation at Light Load
n Current Mode Operation for Excellent Line and Load
Transient Response
n Low Quiescent Current: 270µA
n Shutdown Mode Draws Only 8µA Supply Current
n ±2.5% Reference Accuracy
n Tiny 6-Lead SOT-23 Package
n Lithium-Ion-Powered Applications
n Cellular Telephones
n Wireless Modems
n Portable Computers
n Scanners
L,LT ,LT C ,LT M ,Burst Mode,Linear Technology and the Linear logo are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter
Efficiency vs Load Current
ITH/RUN
LTC1872
147k
422k
80.6k
R1
0.03Ω
L1
4.7µH
220pF
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R226K6.3
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: IRLMS2002
R1: DALE 0.25W
GND
VFB
5
4
6
1872 TA01
1
2
3NGATE
VIN
SENSE–
C1
10µF
10V
VIN
3.3V
VOUT
5V
1A
C2
2×22F
6.3V
+
M1 D1
LOAD CURRENT (mA)
1
EFFICIENCY (%)
100
95
90
85
80
75
70
65
10 100 1000
1872 TA01b
VIN = 3.3V
VOUT = 5V

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pin conFiguraTionabsoluTe MaxiMuM raTings
Input Supply Voltage (VIN)......................... –0.3V to 10V
SENSE–, NGATE Voltages ............. –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages .............................. –0.3V to 2.4V
NGATE Peak Output Current (<10µs) ........................ 1A
Storage Ambient Temperature Range ....–65°C to 150°C
Operating Temperature Range (Note 2)....–40°C to 85°C
Junction Temperature (Note 3) ............................ 150°C
Lead Temperature (Soldering, 10 sec)................... 300°C
(Note 1)
I
TH/RUN 1
GND 2
VFB 3
6 NGATE
5 VIN
4 SENSE
–
TOP VIEW
S6 PACKAGE
6-LEAD PLASTIC SOT-23
TJMAX = 150°C, θJA = 230°C/W
orDer inForMaTion
elecTrical characTerisTics
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1872E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJis calculated from the ambient temperature TAand power
dissipation PDaccording to the following formula:
TJ= TA+ (PD•θJA°C/W)
PARAMETER CONDITIONS MIN TYP MAX UNITS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Typicals at VIN = 4.2V (Note 4)
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V, VITH/RUN = 0V
VIN < UVLO Threshold
270
230
8
6
420
370
22
10
µA
µA
µA
µA
Undervoltage Lockout Threshold VIN Falling
VIN Rising
l1.55
1.85
2.00
2.10
2.35
2.40
V
V
Shutdown Threshold (at ITH/RUN) l0.15 0.35 0.55 V
Start-Up Current Source VITH/RUN = 0V 0.25 0.5 0.85 µA
Regulated Feedback Voltage 0°C to 70°C(Note 5)
–40°C to 85°C(Note 5)
l
l
0.780
0.770
0.800
0.800
0.820
0.830
V
V
VFB Input Current (Note 5) 10 50 nA
Oscillator Frequency VFB = 0.8V 500 550 650 kHz
Gate Drive Rise Time CLOAD = 3000pF 40 ns
Gate Drive Fall Time CLOAD = 3000pF 40 ns
Peak Current Sense Voltage (Note 6) 114 120 mV
The ldenotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA= 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE
LTC1872ES6#PBF LTC1872ES6#TRPBF LTMK 6-Lead Plastic SOT-23 –40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on nonstandard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1872 is tested in a feedback loop that servos VFB to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense
voltage is VREF/6.67 at duty cycle <40%, and decreases as duty cycle
increases due to slope compensation as shown in Figure 2.

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Typical perForMance characTerisTics
Reference Voltage
vs Temperature
Undervoltage Lockout Trip
Voltage vs Temperature
Shutdown Threshold
vs Temperature
Maximum Current Sense Trip
Voltage vs Duty Cycle
Normalized Oscillator Frequency
vs Temperature
TEMPERATURE (°C)
–55
775
VFB VOLTAGE (mV)
780
790
795
800
825
810
–15 25 45 125
1872 G01
785
815
820
805
–35 5 65 85 105
VIN = 4.2V
TEMPERATURE (°C)
–55
–10
NORMALIZED FREQUENCY (%)
–8
–4
–2
0
10
4
–15 25 45 125
1872 G02
–6
6
8
2
–35 5 65 85 105
VIN = 4.2V
TEMPERATURE (°C)
–55
1.84
UVLO TRIP VOLTAGE (V)
1.88
1.96
2.00
2.04
2.24
2.12
–15 25 45 125
1872 G03
1.92
2.16
2.20
2.08
–35 5 65 85 105
VIN FALLING
DUTY CYCLE (%)
20 30
VIN – VSENSE– (mV)
100
1872 G04
40 50 60 70 80 90
130
120
110
100
90
80
70
60
50
VIN = 4.2V
TA= 25°C
TEMPERATURE (°C)
–55
200
ITH/RUN VOLTAGE (mV)
240
320
360
400
600
480
–15 25 45 125
1872 G05
280
520
560
440
–35 5 65 85 105
VIN = 4.2V
pin FuncTions
ITH/RUN (Pin 1): This pin performs two functions.It
serves as the error amplifier compensation point as well
as the run control input.Nominal voltage range for this
pin is 0.7V to 1.9V. Forcing this pin below 0.35V causes
the device to be shut down.In shutdown all functions are
disabled and the NGATE pin is held low.
GND (Pin 2): Ground Pin.
VFB (Pin 3): Receives the feedback voltage froman external
resistive divider across the output.
SENSE–(Pin 4): The Negative Input to the Current Com-
parator.
VIN (Pin 5): Supply Pin.Must be closely decoupled to
GND Pin 2.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VIN.

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FuncTional DiagraM
operaTion
Main Control Loop
The LTC1872 is a constant frequency current mode
switching regulator.During normal operation,the external
N-channel power MOSFET is turned on each cycle by the
oscillator and turned off when the current comparator
(ICMP) resets the RS latch.The peak inductor current
at which ICMP resets the RS latch is controlled by the
voltage on the ITH/RUN pin,which is the output of the
error amplifier EAMP.An external resistive divider con-
nected between VOUT and ground allows the EAMP to
receive an output feedback voltage VFB.When the load
current increases,it causes a slight decrease in VFB
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
–
+
–
+
–
+
0.15V
0.5µA
0.3V
SLEEP
OVP
BURST
CMP
SHDN
1.2V
UV
1872FD
VREF
+
60mV
VREF
0.8V
VIN
RS
VOLTAGE
REFERENCE
SLOPE
COMP
ICMP
R
S
Q
FREQ
FOLDBACK
OSC
SENSE– VIN
5
2
4
–
+
4
EAMP
VFB
–
+
3
NGATE
VIN
6
ITH/RUN
VIN
0.35V
VREF
0.8V
1
–
+
SHDN
CMP
0.3V
GND
–
+
UNDERVOLTAGE
LOCKOUT
VIN
(Refer to Functional Diagram)
relative to the 0.8V reference,which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
The main control loop is shut down by pulling the ITH/
RUN pin low.Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
As the external compensationnetwork continues to charge
up, the corre
sponding output current trip level follows,
allowing normal operation.

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operaTion
Comparator OVP guards against transient overshoots
>7.5% by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Burst Mode Operation
The LTC1872 enters Burst Mode operation at low load
currents. In this mode,the peak current of the inductor is
set as if VITH/RUN = 1V (at low duty cycles)even though
the voltage at the ITH/RUN pin is at a lower value.If the
inductor’s average current is greater than the load require-
ment, the voltage at the ITH/RUN pin will drop.When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET.The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1872 resumes normal operation.The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
Undervoltage Lockout
To prevent operation of the N-channel MOSFET below safe
input voltage levels,an undervoltage lockout is incorpo-
rated into the LTC1872. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
and all circuitry is turned off except the undervoltage block,
which draws only several microamperes.
Overvoltage Protection
The overvoltage comparator in the LTC1872 will turn the
external MOSFET off when the feedback voltage has risen
7.5%above the reference voltage of 0.8V. This comparator
has a typical hysteresis of 20mV.
Slope Compensation and Inductor’s Peak Current
The inductor’s peak current is determined by:
IPK =
V
ITH−
0.7
10 RSENSE
( )
when the LTC1872 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak inductor
current.The amount of reduction is given by the curves
in Figure 2.
Short-Circuit Protection
Since the power switch in a boost converter is not in
series with the power path from input to load,turning off
the switch provides no protection from a short-circuit at
the output.External means such as a fuse in series with
the boost inductor must be employed to handle this fault
condition.
Figure 2. Maximum Output Current vs Duty Cycle
DUTY CYCLE (%)
110
100
90
80
70
60
50
40
30
20
10
SF = IOUT/IOUT(MAX) (%)
1872 F02
0 70 80 90 1006010 20 30 40 50
IRIPPLE = 0.4IPK
AT 5% DUTY CYCLE
IRIPPLE = 0.2IPK
AT 5% DUTY CYCLE
VIN = 4.2V

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The basic LTC1872 application circuit is shown in
Figure1. External component selection is driven by the
load requirement and begins with the selection of L1and
RSENSE (= R1). Next,the power MOSFET and the output
diode D1is selected followed by CIN(= C1) and COUT(= C2).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage de-
veloped across RSENSE,the threshold of the comparator
determines the inductor’s peak current.The output current
the LTC1872 can provide is given by:
IOUT =0.12
RSENSE
−IRIPPLE
2




VIN
VOUT +VD
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section)and VDis the
forward drop of the output diode at the full rated output
current.
A reasonable starting point for setting ripple current is:
IRIPPLE =O.4
( )
IOUT
( )
V
OUT +
V
D
V
IN
Rearranging the above equation, it becomes:
RSENSE =1
10
( )
IOUT
( )
VIN
VOUT +VD





for Duty Cycle <40%
However, for operation that is above 40% duty cycle,slope
compensation’s effect has to be taken into consideration
to select the appropriate value to provide the required
amount of current. Using the scaling factor (SF, in %) in
Figure 2, the value of RSENSE is:
RSENSE =SF
10
( )
IOUT
( )
100
( )
VIN
VOUT +VD





applicaTions inForMaTion
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for thesame amount of inductor ripple
current. However,this is at the expense of efficiency due
to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple
current. The ripple current,IRIPPLE,decreases with higher
inductance or frequency and increases with higher VOUT.
The inductor’s peak-to-peak ripple current is given by:
IRIPPLE =VIN
f L
( )
VOUT +VD−VIN
VOUT +VD





where f is the operating frequency.Accepting larger values
of IRIPPLE allows the use of low inductances,but results
in higher output voltage ripple and greater core losses.
A reasonable starting point for setting ripple current is:
IRIPPLE =0.4 IOUT MAX
( )
( )
VOUT +VD
VIN





In Burst Mode operation,the ripple current is normally set
such that the inductor current is continuous during the
burst periods.Therefore,the peak-to-peak ripple current
must not exceed:
IRIPPLE ≤
0.03
R
SENSE
This implies a minimum inductance of:
LMIN =VIN
f0.03
RSENSE





VOUT +VD−VIN
VOUT +VD





A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.

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Inductor Selection
When selecting the inductor,keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor.Also,keep in mind that
the DC resistance of the inductor will affect the efficiency.
Off the shelf inductors are available from Murata,Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
Power MOSFET Selection
The main selection criteria for the power MOSFET are the
threshold voltage VGS(TH),the “on” resistance RDS(ON),
reverse transfer capacitance CRSS and total gate charge.
Since the LTC1872 is designed for operation down to low
input voltages,a logic level threshold MOSFET (RDS(ON)
guaranteed at VGS = 2.5V) is required for applications
that work close to this voltage.When these MOSFETs are
used, make sure that the input supply to the LTC1872 is
less than the absolute maximum VGS rating,typically 8V.
Therequired minimum RDS(ON)of the MOSFET is governed
by its allowable power dissipation given by:
RDS(ON) ≅
P
P
DC
( )
IIN21+δp
( )
where PPis the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON)
vs temperature curve,but δp = 0.005/°C can be used as
an approximation for low voltage MOSFETs.DC is the
maximum operating duty cycle of the LTC1872.
Output Diode Selection
Under normal load conditions,the average current con-
ducted by the diode in a boost converter is equal to the
output load current:
I
D(avg) =
I
OUT
It is important to adequately specify the diode peak cur-
rent and average power dissipation so as not to exceed
the diode ratings.
Schottky diodes are recommended for low forward drop
and fast switching times.Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
CIN and COUT Selection
To prevent large input voltage ripple,a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
C
IN
Required I
RMS ≈
0.3
( )
I
RIPPLE
where IRIPPLE is as defined in the Inductor Value Calcula-
tion section.
Notethatcapacitor manufacturer’s ripple currentratings are
oftenbased on 2000 hours of life.This makes it advisable to
further derate the capacitor,or to choose a capacitor rated
at a higher temperature than required.Several capacitors
may be paralleled to meet the size or height requirements
in the design.Due to the high operating frequency of the
LTC1872, ceramic capacitors can also be used for CIN.
Always consult the manufacturer if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically,once the ESR require-
ment is satisfied,the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
ΔVOUT ≈IO•VOUT +VD
VIN
+IRIPPLE
2




•
ESR2+1
2πfCOUT








2








1
2

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applicaTions inForMaTion
where f is the operating frequency,COUT is the output
capacitanceand IRIPPLE is the ripple current intheinductor.
Manufacturers such as Nichicon,United Chemicon and
Sanyoshouldbeconsideredforhigh performancethrough-
hole capacitors.The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat higher
price. The output capacitor RMS current is approximately
equal to:
I
PK
•DC−DC2
where IPK is the peak inductor current and DC is the switch
duty cycle.
When using electrolytic output capacitors,if the ripple and
ESR requirements are met,there is likely to be far more
capacitance than required.
In surface mount applications,multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application.Aluminum
electrolytic and dry tantalum capacitors are both available
in surface mount configurations.An excellent choice of
tantalum capacitors is the AVX TPS and KEMET T510
series of surface mount tantalum capacitors.Also,
ceramic capacitors in X5R pr X7R dielectrics offer excel-
lent performance.
Low Supply Operation
Although the LTC1872 can function down to approxi-
mately 2.0V, the maximum allowable output current is
reduced when VIN decreases below 3V. Figure 3shows
the amount of change as the supply is reduced down to
2V. Also shown in Figure 3is the effect of VIN on VREF as
VIN goes below 2.3V.
Setting Output Voltage
The LTC1872 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4).
By selecting resistor R1, a constant current is caused to
flow through R1and R2to set the overall output voltage.
The regulated output voltage is determined by:
VOUT =0.8V 1+R2
R1





INPUT VOLTAGE (V)
2.0
NORMALIZED VOLTAGE (%)
105
100
95
90
85
80
75 2.2 2.4 2.6 2.8
1872 F03
3.0
VREF
VITH
Figure 4. Setting Output Voltage
Figure 3. Line Regulation of VREF and VITH
3
VFB
VOUT
LTC1872
R1
1872 F04
R2

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For most applications,an 80k resistor is suggested for
R1. To prevent stray pickup,locate resistors R1and R2
close to LTC1872.
Efficiency Considerations
The efficiency of a switching regulator isequal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc.are the individual losses as a percent-
age of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1872 circuits:1) LTC1872 DC bias current, 2)
MOSFET gate charge current, 3) I2R losses and 4) voltage
drop of the output diode.
1. The VIN current is the DC supply current,given in the
electrical characteristics,that excludes MOSFET driver
and control currents.VIN current results in a small loss
which increases with VIN.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET.Each
time a MOSFET gate is switched from low to high to
low again,a packet of charge,dQ,moves from VIN
to ground.The resulting dQ/dt is a current out of VIN
which is typically much larger than the contoller’s DC
supply current.In continuous mode,IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of
the MOSFET,inductor and current sense resistor.
The MOSFET RDS(ON) multiplied by duty cycle times
the average output current squared can be summed
with I2R losses in the inductor ESR in series with the
current sense resistor.
4. The output diode is a major source of power loss at high
currents. The diode loss is calculated by multiplying
the forward voltage by the load current.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(VIN)2IIN(MAX)CRSS(f)
Other losses,including CIN and COUT ESR dissipative
losses, and inductor core losses,generally account for
less than 2% total additional loss.

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PC Board Layout Checklist
When laying out the printed circuit board,the following
checklist should be used to ensure proper operation of
the LTC1872. These items are illustrated graphically in
the layout diagram in Figure 5. Check the following in
your layout:
1. The Schottky diode should be closely connected
between the output capacitor and the drain of the
external MOSFET.
2. The (+) plate of CIN should connect to the sense resis-
tor as closely as possible.This capacitor provides AC
current to the inductor.
3. The input decoupling capacitor (0.1µF) should be
connected closely between VIN (Pin 5) and ground
(Pin 2).
4. Connect the end of RSENSE as close to VIN (Pin 5) as
possible. The VIN pin is the SENSE+of the current
comparator.
5. The trace from SENSE–(Pin 4) to the Sense resistor
should be kept short.The trace should connect close
to RSENSE.
6. Keep the switching node NGATE away from sensitive
small signal nodes.
7. The VFB pin should connect directly to the feedback
resistors. The resistive divider R1and R2must be
connected between the (+) plate of COUT and signal
ground.
L1 M1
BOLD LINES INDICATE HIGH CURRENT PATHS
D1
RS
VOUT
VIN
1872 F05
0.1µF
CITH
RITH
R2
R1
CIN
+
ITH/RUN
LTC1872
GND
VFB
6
5
4
1
2
3
NGATE
VIN
SENSE–
COUT
+
Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)