Richtek Rt7285c User manual

RT7285C

DS7285C-03 July 2014 www.richtek.com

Ordering Information

Note :

Richtek products are :

RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

Suitable for use in SnPb or Pb-free soldering processes.

1.5A, 18V, 500kHz ACOTTM Synchronous Step-Down Converter

General Description

The RT7285C is a synchronous step-down converter with

Advanced Constant On-Time (ACOTTM) mode control. The

ACOTTM provides a very fast transient response with few

external components. The low impedance internal

MOSFET supports high efficiency operation with wide input

voltage range from 4.3V to 18V. The proprietary circuit of

the RT7285C enables to support all ceramic capacitors.

The output voltage can be adjusted between 0.6V and 8V.

Features



4.3V to 18V Input Voltage Range



1.5A Output Current



Advanced Constant On-Time Control



Fast Transient Response



Support All Ceramic Capacitors



Up to 95% Efficiency



500kHz Switching Frequency



Adjustable Output Voltage from 0.6V to 8V



Cycle-by-Cycle Current Limit



Input Under-Voltage Lockout



Hiccup Mode Under-Voltage Protection



Thermal Shutdown



RoHS Compliant and Halogen Free

Applications

Industrial and Commercial Low Power Systems

Computer Peripherals

LCD Monitors and TVs

Green Electronics/Appliances

Point of Load Regulation for High-Performance DSPs,

FPGAs, and ASICs

Simplified Application Circuit

Package Type

E : SOT-23-6

J6 : TSOT-23-6

Lead Plating System

G : Green (Halogen Free and Pb Free)

RT7285C

Pin Configurations

(TOP VIEW)

SOT-23-6 / TSOT-23-6

BOOT GND FB

SW VIN EN

Marking Information

0W= : Product Code

DNN : Date Code

0B= : Product Code

DNN : Date Code

RT7285CGE

RT7285CGJ6

VIN

GND

BOOT

SW VOUT

VIN

RT7285C

Enable

0W=DNN

0B=DNN

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Functional Pin Description

Pin No.

SOT-23-6 TSOT-23-6 Pin Name Pin Function

1 1 BOOT

Bootstrap Supply for High-Side Gate Driver. Connect a 0.1F ceramic

capacitor between the BOOT and SW pins.

2 2 GND Power Ground.

3 3 FB

Feedback Voltage Input. The pin is used to set the output voltage of the

converter via a resistive divider. The converter regulates VFB to 0.6V

4 4 EN

Enable Control Input. Connect EN to a logic-high voltage to enable the IC or to

a logic-low voltage to disable. Do not leave this high impedance input

unconnected.

5 5 VIN

Power Input. The input voltage range is from 4.3V to 18V. Must bypass with a

suitable large ceramic capacitor at this pin.

6 6 SW Switch Node. Connect to external L-C filter.

Function Block Diagram

Operation

The RT7285C is a synchronous step-down converter with

advanced constant on-time control mode. Using the ACOT

control mode can reduce the output capacitance and fast

transient response. It can minimize the component size

without additional external compensation network.

Current Protection

The inductor current is monitored via the internal switches

cycle-by-cycle. Once the output voltage drops under UV

threshold, the RT7285C will enter hiccup mode.

UVLO Protection

To protect the chip from operating at insufficient supply

voltage, the UVLO is needed. When the input voltage of

VIN is lower than the UVLO falling threshold voltage, the

device will be lockout.

Thermal Shutdown

When the junction temperature exceeds the OTP

threshold value, the IC will shut down the switching

operation. Once the junction temperature cools down and

is lower than the OTP lower threshold, the converter will

autocratically resume switching.

Comparator

VIBIAS

BOOT

GND

PVCC

Driver

GND SW

Control

UV & OV

Min off

PVCC

Reg

VREF

VIN

On-Time

VIN

PVCC

UGATE

LGATE

Ripple

Gen.

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Parameter Symbol Test Conditions Min Typ Max Unit

Shutdown Current ISHDN V

EN = 0V -- -- 4 A

Quiescent Current IQV

EN = 2V, VFB = 1V -- 0.5 -- mA

Switch-On Resistance High-Side RDS(ON)_H V

BOOTSW = 4.8V -- 230 -- m

Low-Side RDS(ON)_L V

IN = 5V -- 130 --

Current Limit ILIM Valley Current 1.7 2.2 2.8 A

Oscillator Frequency fSW -- 500 -- kHz

Maximum Duty Cycle DMAX -- 90 -- %

Minimum On-Time tON -- 60 -- ns

Feedback Threshold Voltage VFB 591 600 609 mV

EN Input Threshold Logic-High VEN_H 1.5 -- --

Logic-Low VEN_L -- -- 0.4

VIN Under-Voltage Lockout

Threshold VUVLO V

IN Rising 3.55 3.9 4.25 V

VIN Under-Voltage Lockout

Threshold Hysteresis -- 340 -- mV

Electrical Characteristics

(VIN = 12V, TA= 25°C, unless otherwise specified)

Absolute Maximum Ratings (Note 1)

VIN to GND ----------------------------------------------------------------------------------------------------- −0.3V to 20V

SW to GND ---------------------------------------------------------------------------------------------------- −0.3V to (VIN + 0.3V)

< 10ns ----------------------------------------------------------------------------------------------------------- −5V to 25V

BOOT to GND ------------------------------------------------------------------------------------------------- (VSW −0.3V) to (VSW + 6V)

Other Pins------------------------------------------------------------------------------------------------------ −0.3V to 6V

Power Dissipation, PD@ TA= 25°C

SOT-23-6 / TSOT-23-6 --------------------------------------------------------------------------------------- 0.625W

Package Thermal Resistance (Note 2)

SOT-23-6 / TSOT-23-6, θJA --------------------------------------------------------------------------------- 160°C/W

SOT-23-6 / TSOT-23-6, θJC --------------------------------------------------------------------------------- 15°C/W

Lead Temperature (Soldering, 10 sec.) ------------------------------------------------------------------ 260°C

Junction Temperature ---------------------------------------------------------------------------------------- 150°C

Storage Temperature Range ------------------------------------------------------------------------------- −65°C to 150°C

ESD Susceptibility (Note 3)

HBM (Human Body Model) --------------------------------------------------------------------------------- 2kV

Recommended Operating Conditions (Note 4)

Supply Input Voltage, VIN ---------------------------------------------------------------------------------- 4.3V to 18V

Junction Temperature Range ------------------------------------------------------------------------------- −40°C to 125°C

Ambient Temperature Range ------------------------------------------------------------------------------- −40°C to 85°C

RT7285C

DS7285C-03 July 2014www.richtek.com

Note 1. Stresses beyond those listed “Absolute Maximum Ratings”may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θJA is measured at TA= 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. The case

position of θJC is on the top of the package.

Note 3. Devices are ESD sensitive. Handling precaution is recommended.

Note 4. The device is not guaranteed to function outside its operating conditions.

Parameter Symbol Test Conditions Min Typ Max Unit

Soft-Start Time tSS -- 800 -- s

Thermal Shutdown Threshold TSD -- 160 -- C

Thermal Shutdown Hysteresis TSD -- 20 -- C

VOUT Discharge Resistance RDISCHG EN = 0V, VOUT = 0.5V -- 50 100 

UVP Detect 70 75 80

Output Under-Voltage Trip

Threshold Hysteresis -- 10 --

Output Under-Voltage Delay

Time -- 250 -- s

RT7285C

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Typical Application Circuit

VOUT (V) R1 (kΩ) R2 (kΩ)L (μH) COUT (μF) CFF (pF)

5 110 15 10 22 39

3.3 115 25.5 6.8 22 33

2.5 25.5 8.06 4.7 22 NC

1.2 10 10 3.6 22 NC

Table 1. Suggested Component Values

VIN

GND

BOOT

3.6µH

100nF

22µF

10k

VOUT

1.2V

10µF

VIN

4.3V to 18V

RT7285C

CBOOT

CIN

COUT

Enable

CFF

RT7285C

DS7285C-03 July 2014www.richtek.com

Typical Operating Characteristics

Referecnec Voltage vs. Input Voltage

0.590

0.595

0.600

0.605

0.610

4.5 6.5 8.5 10.5 12.5 14.5 16.5 18.5

Input Voltage (V)

Referecnec Voltage (V)

VIN = 4.5V to 18V, VOUT = 1.2V, IOUT = 0A

Switching Frequency vs. Input Voltage

500

520

540

560

580

600

4681012141618

Input Voltage (V)

Switcing Frequency (kHz)1

VOUT = 1.2V, IOUT = 0A

Output Voltage vs. Load Current

1.190

1.194

1.198

1.202

1.206

1.210

1.214

1.218

1.222

1.226

1.230

0 0.3 0.6 0.9 1.2 1.5

Load Current (A)

Output Voltage (V)

VIN = 4.5V to 18V, VOUT = 1.2V

VIN = 18V

VIN = 12V

VIN = 9V

VIN = 5V

VIN = 4.5V

Reference vs. Temperature

0.590

0.595

0.600

0.605

0.610

-50-25 0 25 50 75100125

Temperature (°C)

Reference Voltage (V)

IOUT = 0A

VIN = 12V

Efficiency vs. Load Current

100

0 0.3 0.6 0.9 1.2 1.5

Load Current (A)

Efficiency (%)

VOUT = 1.2V

VIN = 5V

VIN = 9V

VIN = 12V

VIN = 18V

Efficiency vs. Load Current

100

0 0.3 0.6 0.9 1.2 1.5

Load Current (A)

Efficiency (%)

VOUT = 5V

VIN = 12V

VIN = 15V

VIN = 18V

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Time (1μs/Div)

Switching

VOUT

(20mV/Div)

(1A/Div) VIN = 12V, VOUT = 1.2V, IOUT = 1.5A

VSW

(10V/Div)

Time (1μs/Div)

Switching

VIN = 12V, VOUT = 1.2V, IOUT = 0.75A

VOUT

(20mV/Div)

(1A/Div)

VSW

(10V/Div)

Time (100μs/Div)

Load Transient Response

VOUT

(20mV/Div)

IOUT

(1A/Div)

VIN = 12V, VOUT = 1.2V, IOUT = 0.75A to 1.5A

Switching Frequency vs. Temperature

450

470

490

510

530

550

570

590

610

630

650

-50 -25 0 25 50 75 100 125

Temperature (°C)

Switching Frequency (kHz)1

VOUT = 1.2V

VIN = 6V

VIN = 12V

VIN = 18V

VIN = 4.5V

Current Limit vs. Temperature

1.0

1.2

1.4

1.6

1.8

2.0

2.2

2.4

2.6

2.8

3.0

-50 -25 0 25 50 75 100 125

Temperature (°C)

Current Limit (A)

VIN = 6V

VIN = 12V

VIN = 18V

Time (100μs/Div)

Load Transient Response

VOUT

(20mV/Div)

IOUT

(1A/Div)

VIN = 12V, VOUT = 1.2V, IOUT = 0A to 1.5A

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VIN

(10V/Div)

VSW

(10V/Div)

Time (2.5ms/Div)

Power Off from VIN

VIN = 12V, VOUT = 1.2V, IOUT = 1.5A

VOUT

(1V/Div)

ISW

(1A/Div)

Time (5ms/Div)

Power Off from EN

VIN = 12V, VOUT = 1.2V, IOUT = 1.5A

VOUT

(1V/Div)

VEN

(2V/Div)

ISW

(1A/Div)

VSW

(10V/Div)

Time (2.5ms/Div)

Power On from VIN

VIN = 12V, VOUT = 1.2V, IOUT = 1.5A

VOUT

(1V/Div)

VIN

(10V/Div)

ISW

(1A/Div)

VSW

(10V/Div)

Time (2.5ms/Div)

Power On from EN

VIN = 12V, VOUT = 1.2V, IOUT = 1.5A

VOUT

(1V/Div)

VEN

(2V/Div)

ISW

(1A/Div)

VSW

(10V/Div)

RT7285C

DS7285C-03 July 2014 www.richtek.com

Application information

Inductor Selection

Selecting an inductor involves specifying its inductance

and also its required peak current. The exact inductor value

is generally flexible and is ultimately chosen to obtain the

best mix of cost, physical size, and circuit efficiency.

Lower inductor values benefit from reduced size and cost

and they can improve the circuit's transient response, but

they increase the inductor ripple current and output voltage

ripple and reduce the efficiency due to the resulting higher

peak currents. Conversely, higher inductor values increase

efficiency, but the inductor will either be physically larger

or have higher resistance since more turns of wire are

required and transient response will be slower since more

time is required to change current (up or down) in the

inductor. A good compromise between size, efficiency,

and transient response is to use a ripple current (ΔIL) about

20% to 40% of the desired full output load current.

Calculate the approximate inductor value by selecting the

input and output voltages, the switching frequency (fSW),

the maximum output current (IOUT(MAX)) and estimating a

ΔILas some percentage of that current.









OUT IN OUT

IN SW L

VVV

L = Vf I

Once an inductor value is chosen, the ripple current (ΔIL)

is calculated to determine the required peak inductor

current.

















OUT IN OUT

LIN SW

L(PEAK) OUT(MAX)

L(VALLY) OUT(MAX)

VVV

Vf L

I =I 2

Considering the Typical Operating Circuit for 1.2V output

at 1.5A and an input voltage of 12V, using an inductor

ripple of 0.6A (40%), the calculated inductance value is :



1.2 12 1.2

L = = 3.6μH

12 500kHz 0.6





The ripple current was selected at 0.6A and, as long as

we use the calculated 3.6μH inductance, that should be

the actual ripple current amount. The ripple current and

required peak current as below :





1.2 12 1.2

I = = 0.6A

12 500kHz 3.6μH





L(PEAK) 0.6

and I = 1.5A = 1.8A



Inductor saturation current should be chosen over IC's

current limit.

Input Capacitor Selection

The input filter capacitors are needed to smooth out the

switched current drawn from the input power source and

to reduce voltage ripple on the input. The actual

capacitance value is less important than the RMS current

rating (and voltage rating, of course). The RMS input ripple

current (IRMS) is a function of the input voltage, output

voltage, and load current :

OUT IN

RMS OUT(MAX) IN OUT

I =I 1



Ceramic capacitors are most often used because of their

low cost, small size, high RMS current ratings, and robust

surge current capabilities. However, take care when these

capacitors are used at the input of circuits supplied by a

wall adapter or other supply connected through long, thin

wires. Current surges through the inductive wires can

induce ringing at the RT7285C input which could

potentially cause large, damaging voltage spikes at VIN.

If this phenomenon is observed, some bulk input

capacitance may be required. Ceramic capacitors (to meet

the RMS current requirement) can be placed in parallel

with other types such as tantalum, electrolytic, or polymer

(to reduce ringing and overshoot).

Choose capacitors rated at higher temperatures than

required. Several ceramic capacitors may be paralleled to

meet the RMS current, size, and height requirements of

the application. The typical operating circuit use 10μF and

one 0.1μF low ESR ceramic capacitors on the input.

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Output Capacitor Selection

The RT7285C is optimized for ceramic output capacitors

and best performance will be obtained using them. The

total output capacitance value is usually determined by

the desired output voltage ripple level and transient response

requirements for sag (undershoot on positive load steps)

and soar (overshoot on negative load steps).

Output Ripple

Output ripple at the switching frequency is caused by the

inductor current ripple and its effect on the output

capacitor's ESR and stored charge. These two ripple

components are called ESR ripple and capacitive ripple.

Since ceramic capacitors have extremely low ESR and

relatively little capacitance, both components are similar

in amplitude and both should be considered if ripple is

critical.



RIPPLE RIPPLE(ESR) RIPPLE(C)

V =V V



RIPPLE(ESR) L ESR

V =IR





RIPPLE(C) OUT SW

V =8C f

Output Transient Undershoot and Overshoot

In addition to voltage ripple at the switching frequency,

the output capacitor and its ESR also affect the voltage

sag (undershoot) and soar (overshoot) when the load steps

up and down abruptly. The ACOT transient response is

very quick and output transients are usually small.

However, the combination of small ceramic output

capacitors (with little capacitance), low output voltages

(with little stored charge in the output capacitors), and

low duty cycle applications (which require high inductance

to get reasonable ripple currents with high input voltages)

increases the size of voltage variations in response to

very quick load changes. Typically, load changes occur

slowly with respect to the IC's 500kHz switching frequency.

For the Typical Operating Circuit for 1.2V output and an

inductor ripple of 0.46A, with 1 x 22μF output capacitance

each with about 5mΩESR including PCB trace resistance,

the output voltage ripple components are :

RIPPLE(ESR)

V = 0.46A 5m = 2.3mV

RIPPLE(C) 0.46A

V = = 5.227mV

822μF500kHz

RIPPLE

V = 2.3mV 5.227mV = 7.527mV

But some modern digital loads can exhibit nearly

instantaneous load changes and the following section

shows how to calculate the worst-case voltage swings in

response to very fast load steps.

The output voltage transient undershoot and overshoot each

have two components : the voltage steps caused by the

output capacitor's ESR, and the voltage sag and soar due

to the finite output capacitance and the inductor current

slew rate. Use the following formulas to check if the ESR

is low enough (typically not a problem with ceramic

capacitors) and the output capacitance is large enough to

prevent excessive sag and soar on very fast load step

edges, with the chosen inductor value.

The amplitude of the ESR step up or down is a function of

the load step and the ESR of the output capacitor :

VESR _STEP = ΔIOUT x RESR

The amplitude of the capacitive sag is a function of the

load step, the output capacitor value, the inductor value,the

input-to-output voltage differential, and the maximum duty

cycle. The maximum duty cycle during a fast transient is

a function of the on-time and the minimum off-time since

the ACOTTM control scheme will ramp the current using

on-times spaced apart with minimum off-times, which is

as fast as allowed. Calculate the approximate on-time

(neglecting parasitics) and maximum duty cycle for a given

input and output voltage as :



OUT ON

ON MAX

IN SW ON OFF(MIN)

t = and D =

Vf t t

The actual on-time will be slightly longer as the IC

compensates for voltage drops in the circuit, but we can

neglect both of these since the on-time increase

compensates for the voltage losses. Calculate the output

voltage sag as :



OUT

SAG

OUT IN(MIN) MAX OUT

L(I )

V =2C V D V



 

The amplitude of the capacitive soar is a function of the

load step, the output capacitor value, the inductor value

and the output voltage :

OUT

SOAR OUT OUT

L(I )

V =2C V





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Feed-forward Capacitor (Cff)

The RT7285C is optimized for ceramic output capacitors

and for low duty cycle applications. However for high-output

voltages, with high feedback attenuation, the circuit's

response becomes over-damped and transient response

can be slowed. In high-output voltage circuits (VOUT > 3.3V)

transient response is improved by adding a small “feed-

forward”capacitor (Cff) across the upper FB divider resistor

(Figure 1), to increase the circuit's Q and reduce damping

to speed up the transient response without affecting the

steady-state stability of the circuit. Choose a suitable

capacitor value that following below step.

Get the BW the quickest method to do transient

response form no load to full load. Confirm the damping

frequency. The damping frequency is BW.

Figure 1. Cff Capacitor Setting

Cff can be calculated base on below equation :

ff 1

C2 3.1412 R1 BW 0.8



Enable Operation (EN)

For automatic start-up the EN pin can be connected to

VIN, through a 100kΩresistor. Its large hysteresis band

makes EN useful for simple delay and timing circuits. EN

can be externally pulled to VIN by adding a resistor-

capacitor delay (REN and CEN in Figure 2). Calculate the

delay time using EN's internal threshold where switching

operation begins (1.4V, typical).

An external MOSFET can be added to implement digital

control of EN when no system voltage above 2V is available

(Figure 3). In this case, a 100kΩ pull-up resistor, REN, is

connected between VIN and the EN pin. MOSFET Q1 will

be under logic control to pull down the EN pin. To prevent

enabling circuit when VIN is smaller than the VOUT target

value or some other desired voltage level, a resistive voltage

divider can be placed between the input voltage and ground

and connected to EN to create an additional input under

voltage lockout threshold (Figure 4).

Figure 4. Resistor Divider for Lockout Threshold Setting

Figure 2. External Timing Control

Figure 3. Digital Enable Control Circuit

RT7285C

GND

VIN

REN

CEN

RT7285C

GND

100k

VIN

REN

Enable

RT7285C

GND

VIN

REN1

REN2

RT7285C

GND

VOUT

Cff

Internal Soft-Start (SS)

The RT7285C soft-start uses an internal soft-start time

800μs.

Following below equation to get the minimum capacitance

range in order to avoid UV occur.

OUT OUT

LIM

CV0.61.2

T(I Load Current) 0.8

T 800μs







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Output Voltage Setting

Set the desired output voltage using a resistive divider

from the output to ground with the midpoint connected to

FB. The output voltage is set according to the following

equation :

VOUT = 0.6 x (1 + R1 / R2)

Figure 5. Output Voltage Setting

RT7285C

GND

VOUT

For output voltage accuracy, use divider resistors with 1%

or better tolerance.

External BOOT Bootstrap Diode

When the input voltage is lower than 5.5V it is

recommended to add an external bootstrap diode between

VIN (or VINR) and the BOOT pin to improve enhancement

of the internal MOSFET switch and improve efficiency.

The bootstrap diode can be a low cost one such as 1N4148

or BAT54.

External BOOT Capacitor Series Resistance

The internal power MOSFET switch gate driver is

optimized to turn the switch on fast enough for low power

loss and good efficiency, but also slow enough to reduce

EMI. Switch turn-on is when most EMI occurs since VSW

rises rapidly. During switch turn-off, SW is discharged

relatively slowly by the inductor current during the

deadtime between high-side and low-side switch on-times.

In some cases it is desirable to reduce EMI further, at the

expense of some additional power dissipation. The switch



OUT

R2 (V 0.6)

R1 0.6

Place the FB resistors within 5mm of the FB pin. Choose

R2 between 10kΩand 100kΩto minimize power

consumption without excessive noise pick-up and

calculate R1 as follows :

BOOT

0.1µF

RT7285C

Over-Temperature Protection

The RT7285C features an Over-Temperature Protection

(OTP) circuitry to prevent from overheating due to

excessive power dissipation. The OTP will shut down

switching operation when junction temperature exceeds

160°C. Once the junction temperature cools down by

approximately 20°C, the converter will resume operation.

To maintain continuous operation, the maximum junction

temperature should be lower than 125°C.

Under-Voltage Protection

Hiccup Mode

The RT7285C provides Hiccup Mode Under-Voltage

Protection (UVP). When the VFB voltage drops below

0.4V, the UVP function will be triggered to shut down

switching operation. If the UVP condition remains for a

period, the RT7285C will retry automatically. When the

UVP condition is removed, the converter will resume

operation. The UVP is disabled during soft-start period.

Figure 6. External Bootstrap Diode

turn-on can be slowed by placing a small (<47Ω)

resistance between BOOT and the external bootstrap

capacitor. This will slow the high-side switch turn-on and

VSW's rise. To remove the resistor from the capacitor

charging path (avoiding poor enhancement due to

undercharging the BOOT capacitor), use the external diode

shown in figure 6 to charge the BOOT capacitor and place

the resistance between BOOT and the capacitor/diode

connection.

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Thermal Considerations

For continuous operation, do not exceed absolute

maximum junction temperature. The maximum power

dissipation depends on the thermal resistance of the IC

package, PCB layout, rate of surrounding airflow, and

difference between junction and ambient temperature. The

maximum power dissipation can be calculated by the

following formula :

PD(MAX) = (TJ(MAX) −TA) / θJA

where TJ(MAX) is the maximum junction temperature, TA is

the ambient temperature, and θJA is the junction to ambient

thermal resistance.

For recommended operating condition specifications, the

maximum junction temperature is 125°C. The junction to

ambient thermal resistance, θJA, is layout dependent. For

SOT-23-6 / TSOT-23-6 package, the thermal resistance,

θJA, is 160°C/W on a standard four-layer thermal test board.

The maximum power dissipation at TA= 25°C can be

calculated by the following formula :

PD(MAX) = (125°C −25°C) / (160°C/W) = 0.625W for

SOT-23-6 / TSOT-23-6 package

The maximum power dissipation depends on the operating

ambient temperature for fixed TJ(MAX) and thermal

resistance, θJA. The derating curve in Figure 7 allows the

designer to see the effect of rising ambient temperature

on the maximum power dissipation.

Figure 7. Derating Curve of Maximum Power Dissipation

Layout Considerations

For best performance of the RT7285C, the following layout

guidelines must be strictly followed.

Input capacitor must be placed as close to the IC as

possible.

SW should be connected to inductor by wide and short

trace. Keep sensitive components away from this trace.

0.0

0.4

0.8

1.2

1.6

2.0

0 25 50 75 100 125

Ambient Temperature (°C)

Maximum Power Dissipation (W)1

Four-Layer PCB

Figure 8. PCB Layout Guide

BOOT

GND

VIN

VOUT R1 R2

VIN

CIN

CS* RS*

REN

COUT

SW should be connected to inductor by Wide and

short trace. Keep sensitive components away from

this trace. Suggestion layout trace wider for thermal.

Keep sensitive components away

from this trace. Suggestion layout

trace wider for thermal.

Suggestion layout trace

wider for thermal.

The feedback components must be

connected as close to the device as

possible.

The REN component must

be connected to VIN.

Suggestion layout trace

wider for thermal.

Input capacitor must be placed as close

to the IC as possible. Suggestion layout

trace wider for thermal.

VOUT

GND

RT7285C

DS7285C-03 July 2014www.richtek.com

Outline Dimension

AA1

SOT-23-6 Surface Mount Package

Dimensions In Millimeters Dimensions In Inches

Symbol Min Max Min Max

A 0.889 1.295 0.031 0.051

A1 0.000 0.152 0.000 0.006

B 1.397 1.803 0.055 0.071

b 0.250 0.560 0.010 0.022

C 2.591 2.997 0.102 0.118

D 2.692 3.099 0.106 0.122

e 0.838 1.041 0.033 0.041

H 0.080 0.254 0.003 0.010

L 0.300 0.610 0.012 0.024

RT7285C

DS7285C-03 July 2014 www.richtek.com

Richtek Technology Corporation

14F, No. 8, Tai Yuen 1st Street, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should

obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot

assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be

accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third

parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

TSOT-23-6 Surface Mount Package

Dimensions In Millimeters Dimensions In Inches

Symbol Min Max Min Max

A 0.700 1.000 0.028 0.039

A1 0.000 0.100 0.000 0.004

B 1.397 1.803 0.055 0.071

b 0.300 0.559 0.012 0.022

C 2.591 3.000 0.102 0.118

D 2.692 3.099 0.106 0.122

e 0.838 1.041 0.033 0.041

H 0.080 0.254 0.003 0.010

L 0.300 0.610 0.012 0.024