Texas Instruments LM3429 User manual

LM3429
LM3429 LM3429Q1 N-Channel Controller for Constant Current LED Drivers
Literature Number: SNVS616F

LM3429
LM3429Q1
January 19, 2010
N-Channel Controller for Constant Current LED Drivers
General Description
The LM3429 is a versatile high voltage N-channel MosFET
controller for LED drivers . It can be easily configured in buck,
boost, buck-boost and SEPIC topologies. This flexibility,
along with an input voltage rating of 75V, makes the LM3429
ideal for illuminating LEDs in a very diverse, large family of
applications.
Adjustable high-side current sense voltage allows for tight
regulation of the LED current with the highest efficiency pos-
sible. The LM3429 uses Predictive Off-time (PRO) control,
which is a combination of peak current-mode control and a
predictive off-timer. This method of control eases the design
of loop compensation while providing inherent input voltage
feed-forward compensation.
The LM3429 includes a high-voltage startup regulator that
operates over a wide input range of 4.5V to 75V. The internal
PWM controller is designed for adjustable switching frequen-
cies of up to 2.0 MHz, thus enabling compact solutions.
Additional features include analog dimming, PWM dimming,
over-voltage protection, under-voltage lock-out, cycle-by-cy-
cle current limit, and thermal shutdown.
The LM3429 comes in a low profile, thermally efficient TSSOP
EP 14-lead package.
The LM3429Q1 is an Automotive Grade product that is AEC-
Q100 grade 1 qualified
Features
■LM3429Q1 is an Automotive Grade product that is AEC-
Q100 grade 1 qualified (-40°C to +125°C operating
junction temperature)
■VIN range from 4.5V to 75V
■Adjustable current sense voltage
■High-side current sensing
■2Ω, 1A Peak MosFET gate driver
■Input under-voltage protection
■Over-voltage protection
■PWM dimming
■Analog dimming
■Cycle-by-cycle current limit
■Programmable switching frequency
■Thermal Shutdown
Applications
■LED Drivers - Buck, Boost, Buck-Boost, SEPIC
■Indoor and Outdoor SSL
■Automotive
■General Illumination
■Constant-Current Regulators
Typical Boost Application Circuit
300944i9
300944b6
Boost Evaluation Board
9 Series LEDs at 1A
© 2010 National Semiconductor Corporation 300944 www.national.com
LM3429 LM3429Q1 N-Channel Controller for Constant Current LED Drivers

Connection Diagram
30094404
14-Lead TSSOP EP
Ordering Information
Order Number Spec. Package Type NSC Package
Drawing
Supplied As Features
LM3429MH NOPB TSSOP-14 EP MXA14A 94 Units, Rail
LM3429MHX NOPB TSSOP-14 EP MXA14A 2500 Units, Tape and Reel
LM3429Q1MH NOPB TSSOP-14 EP MXA14A 94 Units, Rail AEC-Q100 Grade 1
qualified. Automotive
Grade Production Flow*
LM3429Q1MHX NOPB TSSOP-14 EP MXA14A 2500 Units, Tape and Reel
*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
Pin Descriptions
Pin Name Description Application Information
1VIN Input Voltage Bypass with 100 nF capacitor to AGND as close to the device as possible in
the circuit board layout.
2 COMP Compensation Connect a capacitor to AGND to set compensation.
3 CSH Current Sense High Connect a resistor to AGND to set signal current. For analog dimming, connect
current source or potentiometer to AGND (see Analog Dimming section).
4 RCT Resistor Capacitor Timing Connect a resistor from the switch node and a capacitor to AGND to set the
switching frequency.
5 AGND Analog Ground Connect to PGND through the DAP copper circuit board pad to provide proper
ground return for CSH, COMP, and RCT.
6 OVP Over-Voltage Protection
Connect to a resistor divider from VOto program output over-voltage lockout
(OVLO). Turn-off threshold is 1.24V and hysteresis for turn-on is provided by
20 µA current source.
7 nDIM Not DIM input
Connect a PWM signal for dimming as detailed in the PWM Dimming section
and/or a resistor divider from VIN to program input under-voltage lockout
(UVLO). Turn-on threshold is 1.24V and hysteresis for turn-off is provided by
20 µA current source.
8 NC No Connection Leave open.
9 PGND Power Ground Connect to AGND through DAP copper pad to provide ground return for GATE.
10 GATE Gate Drive Output Connect to the gate of the external NFET.
11 VCC Internal Regulator Output Bypass with a 2.2 µF–3.3 µF, ceramic capacitor to PGND.
12 IS Main Switch Current Sense Connect to the drain of the main N-channel MosFET switch for RDS-ON sensing
or to a sense resistor installed in the source of the same device.
13 HSP LED Current Sense Positive Connect through a series resistor to LED current sense resistor (positive).
14 HSN LED Current Sense Negative Connect through a series resistor to LED current sense resistor (negative).
DAP
(15) DAP Thermal pad on bottom of IC Connect to AGND and PGND. For thermal considerations see (Note 2).
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LM3429 LM3429Q1

Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN, nDIM -0.3V to 76.0V
-1 mA continuous
OVP, HSP, HSN -0.3V to 76.0V
-100 µA continuous
RCT -0.3V to 3.0V
-1 mA to +5 mA continuous
IS -0.3V to 76.0V
-2V for 100 ns
-1 mA continuous
VCC -0.3V to 8.0V
COMP, CSH -0.3V to 6.0V
-200 µA to +200 µA
Continuous
GATE -0.3V to VCC
-2.5V for 100 ns
VCC+2.5V for 100 ns
-1 mA to +1 mA continuous
PGND -0.3V to 0.3V
-2.5V to 2.5V for 100 ns
Maximum Junction
Temperature
Internally Limited
Storage Temperature Range −65°C to +150°C
Maximum Lead Temperature
(Reflow and Solder) (Note 3)
260°C
Continuous Power Dissipation Internally Limited
ESD Susceptibility (Note 4)
Human Body Model 2 kV
Operating Conditions (Note 1)
Operating Junction
Temperature Range −40°C to +125°C
Input Voltage VIN 4.5V to 75V
Electrical Characteristics (Note 1)
Specifications in standard type face are for TJ= 25°C and those with boldface type apply over the full Operating Temperature
Range ( TJ= −40°C to +125°C). Minimum and Maximum limits are guaranteed through test, design, or statistical correlation. Typical
values represent the most likely parametric norm at TJ= +25°C, and are provided for reference purposes only. Unless otherwise
stated the following condition applies: VIN = +14V.
Symbol Parameter Conditions Min
(Note 5)
Typ
(Note 6)
Max
(Note 5)Units
STARTUP REGULATOR (VCC)
VCC-REG VCC Regulation ICC = 0 mA 6.30 6.90 7.35 V
ICC-LIM VCC Current Limit VCC = 0V 20 27 mA
IQQuiescent Current Static 1.6 3.0
VCC-UVLO VCC UVLO Threshold VCC Increasing 4.17 4.50
V
VCC Decreasing 3.70 4.08
VCC-HYS VCC UVLO Hysteresis 0.1
OVER-VOLTAGE PROTECTION (OVP)
VTH-OVP OVP OVLO Threshold OVP Increasing 1.180 1.240 1.280 V
IHYS-OVP OVP Hysteresis Source
Current
OVP Active (high) 10 20 30 µA
ERROR AMPLIFIER
VCSH CSH Reference Voltage With Respect to AGND 1.210 1.235 1.260 V
Error Amplifier Input Bias
Current
-0.6 0 0.6
µA
COMP Sink / Source
Current
10 26 40
Transconductance 100 µA/V
Linear Input Range (Note 7) ±125 mV
Transconductance
Bandwidth
-6dB Unloaded Response
(Note 7)0.5 1.0 MHz
OFF TIMER (RCT)
tOFF-MIN Minimum Off-time RCT = 1V through 1 kΩ35 75 ns
RRCT RCT Reset Pull-down
Resistance
36 120 Ω
VRCT VIN/25 Reference Voltage VIN = 14V 540 565 585 mV
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LM3429 LM3429Q1

Symbol Parameter Conditions Min
(Note 5)
Typ
(Note 6)
Max
(Note 5)Units
PWM COMPARATOR
COMP to PWM Offset 700 800 900 mV
CURRENT LIMIT (IS)
VLIM Current Limit Threshold 215 245 275 mV
VLIM Delay to Output 35 75
ns
tON-MIN Leading Edge Blanking
Time
75 250 450
HIGH SIDE TRANSCONDUCTANCE AMPLIFIER
Input Bias Current 10 µA
Transconductance 20 119 mA/V
Input Offset Current -1.5 01.5 µA
Input Offset Voltage -7 07mV
Transconductance
Bandwidth
ICSH = 100 µA
(Note 7)250 500 kHz
GATE DRIVER (GATE)
RSRC(GATE) GATE Sourcing Resistance GATE = High 2.0 6.0 Ω
RSNK(GATE) GATE Sinking Resistance GATE = Low 1.3 4.5
UNDER-VOLTAGE LOCKOUT and DIM INPUT (nDIM)
VTH-nDIM nDIM / UVLO Threshold 1.180 1.240 1.280 V
IHYS-nDIM nDIM Hysteresis Current 10 20 30 µA
THERMAL SHUTDOWN
TSD Thermal Shutdown
Threshold
(Note 7) 165
°C
THYS Thermal Shutdown
Hysteresis
(Note 7) 25
THERMAL RESISTANCE
θJA Junction to Ambient
(Note 2)
14L TSSOP EP 40 °C/W
θJC Junction to Exposed Pad
(DAP)
14L TSSOP EP 5.5 °C/W
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability
and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in
the Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be
operated beyond such conditions. All voltages are with respect to the potential at the AGND pin, unless otherwise specified.
Note 2: Junction-to-ambient thermal resistance is highly board-layout dependent. The numbers listed in the table are given for a reference layout wherein the
14L TSSOP EP package has its DAP pad populated with 9 vias. In applications where high maximum power dissipation exists, namely driving a large MosFET
at high switching frequency from a high input voltage, special care must be paid to thermal dissipation issues during board design. In high-power dissipation
applications, the maximum ambient temperature may have to be derated. Maximum ambient temperature (TA-MAX) is dependent on the maximum operating
junction temperature (TJ-MAX-OP = 125°C), the maximum power dissipation of the device in the application (PD-MAX), and the junction-to ambient thermal resistance
of the package in the application (θJA), as given by the following equation: TA-MAX = TJ-MAX-OP – (θJA × PD-MAX). In most applications there is little need for the full
power dissipation capability of this advanced package. Under these circumstances, no vias would be required and the thermal resistances would be 104 °C/W
for the 14L TSSOP EP. It is possible to conservatively interpolate between the full via count thermal resistance and the no via count thermal resistance with a
straight line to get a thermal resistance for any number of vias in between these two limits.
Note 3: Refer to National’s packaging website for more detailed information and mounting techniques. http://www.national.com/analog/packaging/
Note 4: Human Body Model, applicable std. JESD22-A114-C.
Note 5: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100%
production tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used
to calculate Average Outgoing Quality Level (AOQL).
Note 6: Typical numbers are at 25°C and represent the most likely norm.
Note 7: These electrical parameters are guaranteed by design, and are not verified by test.
Note 8: The measurements were made using the standard buck-boost evaluation board from AN-1985.
Note 9: The measurements were made using the standard boost evaluation board from AN-1986.
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LM3429 LM3429Q1

Typical Performance Characteristics TA=+25°C and VIN = 14V unless otherwise specified
Boost Efficiency vs. Input Voltage
VO= 32V (9 LEDs) (Note 9)
300944b6
Buck-Boost Efficiency vs. Input Voltage
VO= 20V (6 LEDs) (Note 8)
300944b5
Boost LED Current vs. Input Voltage
VO= 32V (9 LEDs) (Note 9)
300944b8
Buck-boost LED Current vs. Input Voltage
VO= 20V (6 LEDs) (Note 8)
300944b7
Analog Dimming
VO= 20V (6 LEDs) (Note 8)
300944b9
PWM Dimming
VO= 20V (6 LEDs) (Note 8)
300944c0
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LM3429 LM3429Q1

VCSH vs. Junction Temperature
300944b0
VCC vs. Junction Temperature
300944b1
VRCT vs. Junction Temperature
300944b2
VLIM vs. Junction Temperature
300944b3
tON-MIN vs. Junction Temperature
300944b4
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LM3429 LM3429Q1

Block Diagram
30094403
Theory of Operation
The LM3429 is an N-channel MosFET (NFET) controller for
buck, boost and buck-boost current regulators which are ideal
for driving LED loads. The controller has wide input voltage
range allowing for regulation of a variety of LED loads. The
high-side differential current sense, with low adjustable
threshold voltage, provides an excellent method for regulating
output current while maintaining high system efficiency. The
LM3429 uses a Predictive Off-time (PRO) control architecture
that allows the regulator to be operated using minimal exter-
nal control loop compensation, while providing an inherent
cycle-by-cycle current limit. The adjustable current sense
threshold provides the capability to amplitude (analog) dim
the LED current and the output enable/disable function allows
for PWM dimming using no external components. When de-
signing, the maximum attainable LED current is not internally
limited because the LM3429 is a controller. Instead it is a
function of the system operating point, component choices,
and switching frequency allowing the LM3429 to easily pro-
vide constant currents up to 5A. This simple controller con-
tains all the features necessary to implement a high efficiency
versatile LED driver.
CURRENT REGULATORS
Current regulators can be designed to accomplish three basic
functions: buck, boost, and buck-boost. All three topologies
in their most basic form contain a main switching MosFET, a
recirculating diode, an inductor and capacitors. The LM3429
is designed to drive a ground referenced NFET which is per-
fect for a standard boost regulator. Buck and buck-boost
regulators, on the other hand, usually have a high-side switch.
When driving an LED load, a ground referenced load is often
not necessary, therefore a ground referenced switch can be
used to drive a floating load instead. The LM3429 can then
be used to drive all three basic topologies as shown in the
Basic Topology Schematics section.
Looking at the buck-boost design, the basic operation of a
current regulator can be analyzed. During the time that the
NFET (Q1) is turned on (tON), the input voltage source stores
energy in the inductor (L1) while the output capacitor (CO)
provides energy to the LED load. When Q1 is turned off
(tOFF), the re-circulating diode (D1) becomes forward biased
and L1 provides energy to both COand the LED load. Figure
1shows the inductor current (iL(t)) waveform for a regulator
operating in CCM.
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LM3429 LM3429Q1

30094498
FIGURE 1. Ideal CCM Regulator Inductor Current iL(t)
The average output LED current (ILED) is proportional to the
average inductor current (IL) , therefore if ILis tightly con-
trolled, ILED will be well regulated. As the system changes
input voltage or output voltage, the ideal duty cycle (D) is var-
ied to regulate ILand ultimately ILED. For any current regulator,
D is a function of the conversion ratio:
Buck
Boost
Buck-boost
PREDICTIVE OFF-TIME (PRO) CONTROL
PRO control is used by the LM3429 to control ILED. It is a
combination of average peak current control and a one-shot
off-timer that varies with input voltage. The LM3429 uses
peak current control to regulate the average LED current
through an array of HBLEDs. This method of control uses a
series resistor in the LED path to sense LED current and can
use either a series resistor in the MosFET path or the MosFET
RDS-ON for both cycle-by-cycle current limit and input voltage
feed forward. D is indirectly controlled by changes in both
tOFF and tON, which vary depending on the operating point.
Even though the off-time control is quasi-hysteretic, the input
voltage proportionality in the off-timer creates an essentially
constant switching frequency over the entire operating range
for boost and buck-boost topologies. The buck topology can
be designed to give constant ripple over either input voltage
or output voltage, however switching frequency is only con-
stant at a specific operating point .
This type of control minimizes the control loop compensation
necessary in many switching regulators, simplifying the de-
sign process. The averaging mechanism in the peak detec-
tion control loop provides extremely accurate LED current
regulation over the entire operating range.
PRO control was designed to mitigate “current mode
instability” (also called “sub-harmonic oscillation”) found in
standard peak current mode control when operating near or
above 50% duty cycles. When using standard peak current
mode control with a fixed switching frequency, this condition
is present, regardless of the topology. However, using a con-
stant off-time approach, current mode instability cannot oc-
cur, enabling easier design and control.
Predictive off-time advantages:
•There is no current mode instability at any duty cycle.
•Higher duty cycles / voltage transformation ratios are
possible, especially in the boost regulator.
The only disadvantage is that synchronization to an external
reference frequency is generally not available.
SWITCHING FREQUENCY
An external resistor (RT) connected between the RCT pin and
the switch node (where D1, Q1, and L1 connect), in combi-
nation with a capacitor (CT) between the RCT and AGND pins,
sets the off-time (tOFF) as shown in Figure 2. For boost and
buck-boost topologies, the VIN proportionality ensures a vir-
tually constant switching frequency (fSW).
30094499
FIGURE 2. Off-timer Circuitry for Boost and Buck-boost
Regulators
For a buck topology, RTand CTare also used to set tOFF,
however the VIN proportionality will not ensure a constant
switching frequency. Instead, constant ripple operation can
be achieved. Changing the connection of RTin Figure 2 from
VSW to VIN will provide a constant ripple over varying VIN.
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LM3429 LM3429Q1

Adding a PNP transistor as shown in Figure 3 will provide
constant ripple over varying VO.
300944a0
FIGURE 3. Off-timer Circuitry for Buck Regulators
The switching frequency is defined:
Buck (Constant Ripple vs. VIN)
Buck (Constant Ripple vs. VO)
Boost and Buck-boost
For all topologies, the CTcapacitor is recommended to be
1 nF and should be located very close to the LM3429.
AVERAGE LED CURRENT
The LM3429 uses an external current sense resistor (RSNS)
placed in series with the LED load to convert the LED current
(ILED) into a voltage (VSNS) as shown in Figure 4. The HSP
and HSN pins are the inputs to the high-side sense amplifier
which are forced to be equal potential (VHSP=VHSN) through
negative feedback. Because of this, the VSNS voltage is forced
across RHSP to generate the signal current (ICSH) which flows
out of the CSH pin and through the RCSH resistor. The error
amplifier will regulate the CSH pin to 1.24V, therefore ICSH can
be calculated:
This means VSNS will be regulated as follows:
ILED can then be calculated:
The selection of the three resistors (RSNS, RCSH, and RHSP) is
not arbitrary. For matching and noise performance, the sug-
gested signal current ICSH is approximately 100 µA. This
current does not flow in the LEDs and will not affect either the
off state LED current or the regulated LED current. ICSH can
be above or below this value, but the high-side amplifier offset
characteristics may be affected slightly. In addition, to mini-
mize the effect of the high-side amplifier voltage offset on LED
current accuracy, the minimum VSNS is suggested to be
50 mV. Finally, a resistor (RHSN = RHSP) should be placed in
series with the HSN pin to cancel out the effects of the input
bias current (~10 µA) of both inputs of the high-side sense
amplifier. Note that he CSH pin can also be used as a low-
side current sense input regulated to the 1.24V. The high-side
sense amplifier is disabled if HSP and HSN are tied to GND.
30094457
FIGURE 4. LED Current Sense Circuitry
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LM3429 LM3429Q1